High Voltage Supply

The high voltage (HV) supply was one of the first items to be designed and constructed, and has subsequently been modified quite a few times to become what is now a flexible and reliable source of high voltage and current at the line frequency of 50Hz (UK standard), and up to sustained power outputs of 1600W, and peak power outputs up to 2500W.

Note: A high voltage supply is capable of delivering voltages and currents, even at lower powers, that are instantly lethal, and that any design and operation of a high voltage unit should be undertaken with great care by a trained and experienced individual. In my own case I was trained to work with high voltage equipment early in my career as an electronic engineer, and hence have opted for experimental flexibility, and maximum configuration, a power supply build that is accessible, open, and where high voltages could externally be exposed to the operator at certain key points. Careful design, implementation, and operation of such a supply is for the full responsibility of the individual concerned.

In the early days of this research it was unclear to me where in the experimental circuit interesting and unusual electrical phenomena originated from, whether it was the product of the generator, the tuning and driving units, the experimental coils themselves, the driven loads, the surrounding environment, or a combination of these factors. Later in the research I discovered that the generation of particularly displacement related events required a number of pre-conditions to be established, which involved the balance of the electric and magnetic fields of induction, which in turn involves all of the above factors, combined with a non-linear trigger, and with a defined load or “need” that cannot be met through the process of transference. These pre-conditions and the details pertaining to the generation of a displacement event will be considered and written-up in subsequent posts.

It was considered central to the early research, in replicating the key measurements and observations of other significant works e.g. Dollard et al[1], that the generator design be as close as possible to those used, and especially considering that actual units and components may not be easily obtainable e.g. an original H.G. Fischer diathermy unit. However for this unit certain videos, internal pictures, and schematics where obtainable online and formed the basis of the first stage of building a suitable generator using easily obtainable parts and components. The overall generator to be used for experimentation is a combination of the HV supply detailed in this post, and driving a range of subsequent generator stages that transform the supplied HV AC voltage and currents at the line frequency, into higher frequency ac, oscillations, impulses, bursts, modulated waveforms, and other such driving waveforms as may be useful to the study of the displacement and transference of electric power.

Figures 1 show the HV supply which currently drives the different types of generator stages:

The circuit diagram for the HV supply is shown in Figure 2 below, or click here to view the high resolution version. The schematic should be referenced for the subsequent circuit description:

AC line power at UK standard 240V 50Hz is fed via a high current (16A) 3-pin connector to a domestic distribution box with 3 circuit breakers at 6A, 10A, and 20A. The 6A circuit breaker feeds a low voltage switched-mode power supply unit providing 15V @ 3A and is used to power low voltage circuits in the HV supply and the generator stage including, pre-amplifiers, fans, control electronics, measurement devices, indicators, meters, and any other low voltage units. The HV supply is arranged with a number of low voltage two-pin power jack sockets to supply low voltage generator stage requirements on the upper level. The 10A circuit breaker powers filament transformers for vacuum tube generator stages, and auxiliary devices requiring line voltage AC, via a suitable mains output connector in the form of shielded 3-pin connector, and ceramic connection block for ad-hoc connections. The 20A circuit breaker feeds the high voltage transformers via a suitable power controller.

The distribution unit also has space for an incoming line RCD breaker, but has subsequently been removed, as it was found to be too sensitive to some experiments where power is reflected back into the HV supply, and causing the RCD to cut the power during the experiment. As an alternative a larger RCD was incorporated into the mains distribution for the laboratory, and separate mains circuits fed via a UPS (uninterruptible power supply) to measurement and test equipment, and computers. This arrangement prevents sensitive equipment from inadvertently being switched-off and/or rebooted during certain experiments when the lab RCD would disengage to protect the input supply. Having the test equipment running under these circumstances has proved key to understanding conditions and events within the experiment that have caused large reflections back through the mains supply.

The 20A circuit is first passed through a high current line filter which is used to prevent higher frequency electrical disturbances from being reflected back into the mains supply, and offer a measure of isolation between the two. The output of the line filter is fed to a power controller which enables the variable control of power supplied to the high voltage transformers. This HV supply was specifically designed around the use of the microwave oven transformer (MOT) as the high voltage part of the supply. MOTs are very readily available, and have proved to be a strong and robust transformer for this type of supply. The transformers used are all Galanz GAL-900E based which nominally produce 2100VRMS @ 900W ~0.45ARMS, and are quite common in UK domestic microwave ovens. The MOT represents a significant inductive load to the incoming AC supply, and uncorrected will reduce the power factor from the ideal 1 to ~0.6. To correct for this and reduce the draw on the incoming supply a power factor correction (PFC) capacitor can be used at the AC line input (after the line filter and before the SCR). A 20µF AC PFC capacitor can be used in order to correct for a single MOT. For 2-3 MOTs being used together this can be increased to 40µF.

The MOT is a transformer designed to drive a specific impedance load, (magnetron via a voltage doubler and tuned with a series capacitor), with the minimum quantity, and hence cost, of copper, and with the cheapest and simplest manufacture methods and components. This leads to certain drawbacks in the transformer characteristics, and most especially saturation of the transformer core when driven open-circuit, or connected to a higher than intended load impedance. The core is cheaply manufactured from steel laminate and then welded together which shorts the laminates out, greatly increasing the core saturation rate when adequate power is not drawn from the transformer. A detailed study of the characteristics of the MOT has been presented by Wokoun[2].

The easy core saturation requires the current to be restricted in the primary coil. This is not easily done directly with a variac, as is usual for variable output control of a transformer, since the core easily saturates at low input primary voltages leading to large run-away currents in the primary, rapid core heating, and ultimately destruction of the transformer from excessive heating, not to mention the dangerous risk of a transformer fire which is very hard to deal with due to extreme heating of the steel core even after the power has been cut-off at the input. Instead the current must be restricted either via an inductive load in the primary circuit, or much better, an SCR power controller.

A suitable simple series inductor is the primary of another MOT, (with the secondary shorted), connected in series with the primary of the MOT to act as the HV transformer. Alternatively the secondary of another MOT, (with the primary shorted), can be connected in series with the secondary of the HV transformer MOT, but more heating tends to occur in this configuration from the higher secondary impedance. The series primary connected MOT was found to limit the output current to a degree, and made adjustment with an input variac possible, with less chance of core saturation, but with however limited overall range of adjustment and suitability to changing load impedance. The advantage however of this first method is that the output of the MOT (not in core saturation) is a complete sine wave. In core saturation the output becomes progressively distorted towards a heavily clipped sine wave. It was concluded that this method of power control would be too limited for the wide range of generators that the HV supply would be driving.

The second and preferred method is the SCR based power controller, (similar to a light-dimmer controller but more powerful), which controls the on part of the sinusoidal cycle, and hence controls the overall power delivered to the transformer, which effectively restricts the core saturation whilst providing variable control of the output power. Suitable SCRs are very easily and cheaply available, and a complete unit with an output power of up to 3kW has been used in the HV supply. The disadvantage of the SCR is that the output is no longer a sine wave, but rather a distorted waveform that represents a small part of the total cycle. This has however provided some unexpected benefit in burst and impulse modes that will be discussed in the generator posts, but suffice to say here that the fast SCR turn-off can create very large voltage spikes in the MOT primary as the field collapses, which in turn produces strong impulses in the secondary at the line frequency. These impulses in the secondary, when fed directly to the experiment without a capacitor tank circuit, acts as one method of generating a non-linear trigger for a displacement event.

When working directly with the experiment at hand it is not convenient to keep walking backwards and forwards to the power supply to adjust power level or turn on or off the supply. To enable more distant control of the SCR the variable resistor used to control the power level was removed from the SCR circuit, and positioned in a small plastic box along with an on-off switch. The control box was then attached to the SCR via a long two-wire mains lead (5m), where on-off function is created by switching a higher resistance into the two-wire line, and hence holding the SCR in the off condition, which is also the case if the remote box is disconnected from the SCR power controller. Power control is affected from the variable resistor by reducing the resistance from 500kΩ down to 0Ω, which progressively turns the SCR on for a proportion of the ac line cycle.

Figures 3 below show waveforms from the HV supply at a range of different points in the high voltage supply, and including the high voltage rectifier and tank capacitor at the output to form a load.

The output of the SCR power controller is passed through a system of connections to allow the MOTs to either be driven directly from an external source as required, or by direct connection to the SCR. Each individual MOT can be switched independently to the SCR output allowing the transformers to be used individually or combined in parallel or series combinations to increase the available output current or voltage. The output of the SCR is also fed to a 25W mains incandescent lamp which indicates clearly to the operator when voltage is applied to the input of the one or more of the MOTs. This is a simple but important safety factor when working in the experimental environment, and is a rapid but not exhaustive check to the running status of the high voltage supply. It must also always be remembered that considerable energy can be stored in the generator components, such as tank capacitors etc., and that a no visible lamp output is not a direct indication that it is safe to touch any part of the high voltage circuits prior to the appropriate discharge procedures.

The MOT is a cheaply manufactured component with minimum materials and quality, and hence the high voltage winding isolation to the steel core will not usually withstand voltages in excess of ~1.5 times the nominal designed output. This makes it difficult to combine MOTs in series where the core connected terminal of the secondary has been detached from the core in order to float the secondary, whilst keeping the steel core connected to earth for safety purposes. In this configuration the open-circuit peak voltage of the secondary can reach almost ~6kV from 2 series connected MOTs, which can easily arc-over to the steel core through the secondary insulation. When allowed to happen for any period of time the secondary coil is easily permanently damaged.

To overcome this problem and to enable two MOTs to be connected safely in series (both cores earthed), the MOTs are connected in series anti-phase, or center-tapped arrangement. In this configuration the two cores are connected together to earth, which also means the two core connected ends of the two secondary coils are also connected to earth. The primaries of the two transformers are then connected in reverse phase to each other, (as shown in the circuit diagram), such that one transformer produces +VHT out, and the other transformer produces -VHT out. The total output voltage of the series connected secondaries is 2VHT, and the maximum secondary to core voltage on either transformer is only VHT, preventing any secondary to core breakdown.

Of the three available MOTs in the high voltage supply, two are centre-tap connected, and one is floated from the core. This combination was found to be most flexible where the centre-tapped pair are suitable for driving spark gap based generators, and the floated individual is most suitable for driving vacuum tube based generators and if required in conjunction with a diode voltage doubler. In some generator configurations it was necessary to reduce the secondary current using a power resistor, which also in some specific cases helps to stabilise changes in power factor when driving varying or fluctuating high impedance loads from the generator outputs. When and where required a fan-cooled 100Ω 100W wire-wound resistor was used to reduce secondary currents and stabilise the supply output impedance to the next stage. For sustained outputs the MOTs and output components are cooled using a pair of low voltage fans which are manually switched as required.

Overall the high voltage supply has proved to be robust and versatile in providing high voltage in a variety of configurations to a range of different types of generator circuits. The design of the high voltage supply makes it easy to use in the experiments, with accurate and remote control of the output, and constructed with basic and readily available components.

Click here to continue to the 811A vacuum tube generator.


1. Dollard, E. & Lindemann, P. & Brown, T., Tesla’s Longitudinal Electricity, Borderland Sciences Video, 1987.

2. Wokoun, P., Investigations on Using a Salvaged Microwave Oven Transformer, 2003, KH6GRT Website

 

Vacuum Tube Generator (811A) – Part1

The vacuum tube generator (VTG) mainly used for experiments in the displacement and transference of electric power is based around a pair of 811A power triode vacuum tubes, of either RCA or Russian origin, and with electrical characteristics generally as defined in the RCA 811A datasheet. The 811A’s have demonstrated to be highly flexible, with high reliability, and good overall medium power performance, versus cost and availability, when used in a variety of different configurations. The final output of this generator with these tubes can provide a maximum sustained RF output power of ~600W, and peak output for short bursts (up to 10s) of 900W, and over a wide frequency band up to ~5Mc/s. In addition, the design and implementation of this generator has been arranged in such a way that it can be used in a variety of different configurations, including:

1. A tuned plate class-C Armstrong oscillator which derives automatic feedback from a pick-up coil placed close to the secondary coil.

2. A variable frequency Hartley power oscillator when combined with an independent oscillator drive module.

3. The output stage of a linear power amplifier, when fed with a suitable drive waveform in grounded grid, grid biased, or cathode follower configurations.

4. A modulator stage with suitable cathode keying via a mechanical or semiconductor switching circuit.

5. CW, burst, and modulated oscillation modes when used with or without a dc smoothing capacitor in the plate HV supply.

Other VTGs using the RCA 833A/C and Eimac 4-400A/C are more suited to higher power experiments including, telluric transmission and strong non-linear impulses for displacement experiments, and will be reported in subsequent posts.

Figures 2 below shows an overview of the VTG apparatus, complete with the high voltage supply, the independent Hartley power oscillator module (HPO), the high voltage bridge rectifier (HVBR), and the supply dc smoothing capacitor.

The circuit diagram for the VTG and peripherals is shown in Figure 2 below, or click here to view the high-resolution version.

The high voltage supply VHT (described here), is connected to the input of the HVBR. The rectifier consists of 4 sets of 2 x HVP20 20kV 750mA high voltage diodes in parallel and arranged in a bridge configuration. The output terminal of the bridge B is connected to a tank capacitor 60µF 4kV k75-40a Russian pulse capacitor which results in a smoothed dc output suitable for rapid charging and discharging through a primary, or as a stabilised dc supply for the VTG plate circuit.

A dc meter is also so arranged on the output B to read from 0-4kV in configurable steps based on where the base terminal of the meter is connected to a resistor divider. The high voltage dc meter is a simple arrangement using a 1mA fsd (full-scale deflection) ammeter connected to a resistor divider with 4 x 1MΩ resistors. With all four resistors in series 4kV will result in a 1mA passing through the meter moving it to fsd, with one resistor in series 1kV will result in meter fsd, and accordingly for the 2 and 3kV scales. The meter face has been recalibrated to indicate needle reading in kV.

Before considering each of the configurations listed above in detail, it is necessary to cover the basics of the VTG design. The plate supply is provided by the HVBR output B+ and is a variable 0-4kV peak supply either smoothed or unsmoothed based on whether the rectifier tank capacitor is connected or not. The plate supply B+ is connected directly to the driven load LPCP, which in this case is the primary coil in parallel with the primary tuning capacitor. In this arrangement the primary load (parallel resonant circuit) would present the highest impedance to the vacuum tubes when at resonance. When the primary capacitor is so adjusted to set the secondary resonance frequency it is optimal for the impedance of the primary load to be equal to the output impedance of the dual 811A vacuum tubes in parallel. In this way maximum power is developed in the primary load or through the primary of the coil.

It is important to note that when using a VTG the primary circuit, (in this case the load LPCP), is NOT arranged to resonate at the same frequency as the secondary circuit, which avoids large primary currents being developed in the VTG which would lead to excess plate dissipation in the vacuum tubes, and rapid degrading or destruction of the tubes. Rather the impedance of the primary load at the resonant frequency of the secondary coil defines the driving impedance presented to the vacuum tubes, which in turn are adjusted by the grid bias or feedback to match their internal impedance to the driving impedance, leading to maximum power being developed in the primary circuit whilst not overloading the vacuum tubes beyond their maximum ratings.

Hence the VTG drives the coil in a linear sinusoidal CW mode with optimally arranged load impedance matching, as compared with for example, a spark gap generator where the primary and secondary resonance frequencies are arranged to be equal to allow maximum power transfer from the maximised primary currents during the ring-down phase of the primary tank discharge.

The load of the primary is connected to the plates of the vacuum tubes via high frequency chokes L1R1 and L2R2. These chokes quench very high frequency oscillations which can be generated by discharges within the vacuum tubes during overload conditions, and in turn help prevent very high frequency oscillation run-away conditions which can lead to rapid tube destruction.

The cathodes of the vacuum tubes are bridged by RF bypass capacitors to minimise the impedance of the high frequency signal path, and then connected via a sequence of jumpers J1-J3 to allow for different configuration modes. The cathodes are further connected to the heater power supply which provides a constant bias to the cathode combined heater element of the 811A. The heater power supply is a mains step-down transformer so arranged to provide an ac supply of 6.3VRMS @ 8ARMS for two 811A heater circuits in parallel. The heater supply transformer is fine adjusted in this case by R5 which reduces the voltage across the heater elements and has an RF bypass capacitor C5 to again minimise the impedance of the high frequency signal path. The ac voltage across the tube heater elements is monitored using a calibrated 0-10VRMS ac voltmeter, and R5 adjusted, (in this case 4 x 0R1 25W series connected resistors), to provide an optimal 6.3VRMS +- 5%.

When the vacuum tube generator is used as a linear power amplifier in grounded grid (cathode driven) configuration it is necessary to prevent the RF input signal from feeding back into the heater supply and being dissipated in the low impedance power supply stage. To prevent this bifilar high frequency chokes can be used between the tube cathodes and the heater power supply. At low-frequency, (50Hz for the heater supply), current can pass through the choke from the heater supply to the tube cathodes, but the RF signal at the cathode is prevented from feeding back into the heater supply. Modulation and switching via cathode keying can also be arranged via mechanical or semiconductor switches, and allows for a range of “switched” experiments important in the comparisons of electrical phenomena experienced in the exploration of the displacement and transference of electric power.

The grid of the VTG is arranged with jumpers J4-J6 to allow for different configuration modes, and a current limit resistor R3 to restrict the maximum grid current below the nominally rated for the 811A, and according to the configuration it is used in. In addition both 811A’s are forced cooled by the fan driven by the auxiliary 15V supply. Force cooling allows for higher sustained power in CW modes, and affords an additional protection during peak power overloads. It is quite normal for the 811A plate to glow slightly red under higher sustained powers, and in peak power for short periods 0-10s for it to glow intensely red. Sustained high peak powers > ~650W will lead to plate dissipation overload combined with flash-overs between the plate and grid causing rapid grid damage. In most experiments in the displacement and transference of electric power I have found a comfortable sustained power between 400-600W, which leads to long vacuum tube life, and well sustained tube characteristics according to the nominal data.

Construction of the VTG and HVBR modules are shown in detail in Figures 3 below. For ease and simplicity of experimentation the VTG is open assembled on a simple board with simple insulated mountings and a combination of metal, plastic, and wooden mounts for the various components. Whilst this does lead to a very quick and flexibly modifiable prototype generator, there are significant EMC, interference, temperature, and stability benefits to housing the entire VTG build within a screened metal case, with directed cooling inlet and outlets, and with careful consideration to connection of high current and tension paths with minimal inductance copper bars etc.

Figures 4 show the construction of the Hartley power oscillator module which is considered further in configuration option 2 below.

The different configurations of the VTG module are now considered in more detail:

1. Tuned plate class-C Armstrong oscillator

This configuration is very well suited to investigations at the upper and lower resonant frequency of the coil being driven (FU and FL). In this case the VTG is a series-fed version of the Armstrong oscillator deriving the grid bias feedback from a pickup coil placed in proximity to the secondary coil. In the case of the flat coil this is a small 10 turn cylindrical coil (diameter 100mm) mounted behind the secondary coil and on axis with the coil centres. Oscillation via the pickup coil feedback automatically keeps the VTG on the resonant frequency being explored and adjusts automatically to maintain the resonant frequency when loading is applied to the secondary coil outputs.

In this configuration J1-J3 are left open and J4 is connected. Circuit operation is as follows. When first turned on the impulse current from the tank circuit B+ conducting through the vacuum tube causes a ringing oscillation (ping) in the primary circuit LpCp. This oscillation couples to the secondary circuit LsCs which is further coupled by the pickup coil to the grid bias leakage circuit. When the phasing of the pickup coil is the correct way round for positive feedback, the grid bias leakage capacitor C1 becomes negatively charged during the positive half cycle in the primary circuit, pushing the grid voltage down and progressively restricting conduction in the vacuum tubes towards the off state with much reduced plate current. The negatively charged grid leakage capacitor C1 then discharges through the grid leakage resistor R4. As this happens the grid voltage on the vacuum tube starts to rise progressively towards 0 volts turning on the vacuum tubes with an increasing plate current. The plate current through the primary circuit LpCp again is coupled to the secondary LsCs and the cycle repeats. With the grid bias leakage circuit correctly adjusted the VTG will oscillate with a linear sinusoidal output optimised for maximum plate voltage and current swing, (maximum power transfer at the resonant frequency of the secondary), whilst keeping the grid bias currents within the maximum ratings for the vacuum tubes used.

When setting up this mode of operation it is most important to ensure correct phasing of the pickup coil in connection to the grid bias leakage circuit, and that the values of C1 and R4 are suitably adjusted to allow maximum swing of the plate circuit whilst keeping the grid bias within the maximum ratings. For this VTG with 2 x 811A vacuum tubes R4 is optimally between 1-1.5kΩ and C1 between 1.5-3nF. Considerable power dissipation occurs in R4 when running the VTG at higher powers > 400W, hence the need for a wire-wound power resistor (100W), and preferably as part of the forced cooled air circulation. Excessive or too little grid bias current will lead to a distorted and clipped oscillation, or in extreme cases, no oscillation at all and rapid vacuum tube degradation or destruction. Initial setting up is best done with low plate voltage ~ 500V and higher values of R4. R4 can then be progressively reduced to increase grid bias whilst ensuring a clean and stable output oscillation.

As seen in previous impedance measurement posts for the flat coil (1S-3P) the resonant frequency of the secondary can be adjusted by changing the capacitance of the variable vacuum capacitor in the primary Cp. When adjusted the oscillating frequency of the VTG automatically changes to track the changes in the secondary resonant frequency. At high values of Cp > ~650pF the impedance of the lower resonant frequency FL is dominant and the final oscillating frequency can be adjusted around this frequency FL in the range 1.5-2.2Mc/s. At low values of Cp < ~650pF the impedance of the upper resonant frequency FU dominates  and the final oscillating frequency is in the range 2.7-3.8Mc/s.

Adjusted in this fashion this configuration of the VTG is very well suited to continuous linear measurements for single wire currents, displacement and transference of electric power, and telluric transmission experiments. This configuration is best suited to exploring frequency regions centered around FU and FL, where oscillation is stable and conduction currents in the secondary of the experimental coil are > 10mARMS. This configuration is not suitable for exploring frequency regions far from resonance, at very low bias currents, or between the transition between FU and FL. In these cases it is necessary to use the VTG in modes 2 or 3 where accurate progressive frequency control is provided by an external source and the VTG acts as a power stage, whether that be as a tuned power oscillator, or as a linear power amplifier.

2. Variable frequency Hartley power oscillator

The Hartley power oscillator converts the VTG to a linear oscillator which can be adjusted for variable frequency in bands defined by the combination of band capacitors connected on the HPO module. The frequency of oscillation of the VTG is now determined by the resonant circuit formed on the HPO board, the pickup coil of configuration 1 is not connected in this arrangement. This configuration is suitable for measurements across the entire frequency band of interest, in the case of the flat coil 1S-3P between 1.5-3.5Mc/s.

The HPO must be manually tuned or retuned to a specific frequency of interest and particularly when changing the loading of the secondary coil circuit. The resonant frequency of the secondary coil is very sensitive to electrical loading, temperature, material losses, changing boundary conditions and proximity, which all cause deviations of the configured frequency. Any change in this tuned frequency  requires re-adjustment making the HPO configuration not well suited to experiments designed to explore different loads and operating conditions. It is however very suited to exploring circuit operation where the loading conditions are relatively fixed, and particularly in off-resonance, low current, and frequency transition regions.

In this configuration J1-J3 are left open and J5 is connected to the Hartley oscillator module along with the plate voltage and RF ground as shown on the circuit schematic of Fig 2. The bands of frequency can be adjusted by hardwired jumpers on the HPO module, where the static band capacitors are combined with the 1000pF variable capacitor, and form a parallel resonant circuit with L5 the HPO coil with an inductance of 3.0uH. The available bands are broadly as follows (static combinations of capacitors shown only):

a. 1 x 500pF = 2.4 – 4.1Mc/s

b. 1 x 1000pF = 2.1 – 2.9Mc/s

c. 1 x 500pF + 1 x 1000pF = 1.8 – 2.4Mc/s

d. 2 x 1000pF = 1.7 – 2.1Mc/s

e. 1 x 500pF + 2 x 1000pF = 1.5 – 1.8Mc/s

The setup of the HPO again requires a balance between largest plate swing (output power) without distorting the output wave, whilst restricting the grid bias within maximum parameters. The required frequency band is first configured using the hardwired jumpers, and the HPO grid bias variable resistor is set in the higher halve of its resistance range . The VTG plate supply is first set low at 500V and the HPO coil tapping point starts close to the RF ground end. The tapping point can be progressively moved upwards towards the plate voltage end of the coil until a point is reached where the output of the VTG is stable, clean, and with good power output as the plate supply is progressively increased up to the maximum ratings for the vacuum tubes. The HPO grid bias resistor can then be progressively reduced keeping the grid bias current within the maximum recommended. If no oscillation can be obtained the grid bias resistor can be progressively reduced until oscillation starts, whereupon the other adjustments discussed can be continued with.

The HPO can then be varied across its band frequency range according to the experimental requirements of the circuit, and the output power adjusted using the plate supply voltage. If the loading on the secondary coil changes significantly the HPO will need to be re-tuned and/or re-adjusted to provide stable power oscillation. Off resonance of the secondary coil in the experiment may require the HPO tapping point to be re-adjusted towards the RF ground to prevent excessive power dissipation in the coil. During continuous operation the temperature of the HPO inductor coil should be monitored in order to prevent over-heating. Re-adjustment will be required when moving often and rapidly between on-resonance and off-resonance operating points. As previously stated this configuration mode is best suited to exploration of operating points not accessible with configuration 1, and where the operating conditions do not change rapidly during the measurement cycle.

3. Linear amplifier output stage

The VTG can be operated as the power stage of a linear amplifier when correctly configured and connected to a suitable frequency generator with power amplifier output stage. No pre-amplifier stage is currently provided in the VTG so the driving signal source will need to generate an output between ~ 1-10W in order to the drive the VTG output to a usable output power. In this configuration the VTG can be driven in grounded grid, grid biased, or even as a cathode follower with some change of circuit connection and setup.

For grounded grid operation J1 is connected to the external signal source, and J6 is connected directly to RF ground. For grid driven mode J1-J3 are open and J5 is connected to the external signal source which in this case also needs to arrange for the driving signal to provide suitable dc grid biasing for the vacuum tubes used. For this reason grounded grid operation is preferred for simple linear amplifier operation.

When correctly setup and driven the VTG in this configuration can provide a very stable oscillation output which can be easily and finely adjusted by the external signal source, overcoming some of the adjustment problems of the HPO, but not exceeding the very good power output levels of the HPO. It has been found that the HPO is better when higher power experimentation is required, but the linear amplifier is better for overall signal stability and accurate adjustment.  This configuration also allows for non-sinusoidal waveforms to be applied to the experimental system.

4. Modulator stage

Modulation of the VTG is currently by cathode keying, allowing conduction through the vacuum tubes to be switched at low frequencies via mechanical switches and relays, or at higher frequencies < 100kc/s  by semiconductor MOSFET switches. Modulation of this kind was originally considered to be important in the exploration of the displacement of electric power where non-linear events play a very significant part in the unusual electrical phenomena observed within an electrical system. It has since been found through experimentation that the non-linear impulses generated by this modulation method are not of sufficiently low transition time and low pulse width to be particularly useful in the generation, observation, and measurement of displacement phenomena.

5. CW and burst oscillation modes

The VTG can be arranged in any of the configuration modes 1-4 and then operated in CW or burst modes simply by removing the plate supply tank capacitor at B+. When the tank capacitor is connected at the output of the HVBR a constant plate supply voltage B+ is applied to the primary load. In this case the output of the VTG in oscillation will be in CW mode providing a constant and continuous oscillation wave to the primary circuit LpCp.

When the tank capacitor is removed from the HVBR the output is an unsmoothed full wave rectified supply at 100c/s (based on UK line frequency). Applied to the VTG this produces bursts of oscillation inside the supply envelope, and has proven to be useful in establishing certain operating conditions in experiments orientated to displacement of electric power. These operating conditions and experiments are currently work in progress and will be reported in subsequent posts.


Overall the VTG has proven itself to be a versatile and reliable linear power source suitable to drive a wide range of experiments to explore the displacement and transference of electric power, and also some preliminary lower power telluric transmission experiments. Over several years of operation only 1 x 811A has been replaced after grid flash over destroyed one of the tubes when being used in configuration 2 with the HPO at an off-resonance operating point, with large output mis-match and high reflected power. The VTG has also sustained reliably through high power experiments where both plates of the tubes are glowing bright red for short periods of time. The 811A in RCA and Russian forms have proven themselves to be robust and reliable provided the heater element (spring tensioned) has not broken during extended storage. VTGs with other vacuum tube types and configurations will be presented in subsequent posts.

Click here to continue to part 2 of the 811A vacuum tube generator where the operating characteristics for each of the defined configurations are measured, optimised, and tuned.

 

1920s H.G. Fischer Diathermy

Later in the research, (and after the replica diathermy unit had been designed and built), I was lucky enough to come across a real 1920s H.G. Fischer diathermy unit (HGF), which although being sold untested, and in unknown condition, looked suprisingly good from the pictures. It survived the shipping from the USA to the UK all in one piece, and on closer inspection proved to be in good physical and working condition, including the thermo-ammeter, and the original fuse. The only part missing was the 6V power indicator bulb.

The unit is an early model GP, which is actually a model G in a compact portable cabinet, with a robust carry handle, and metal cabinet corner protectors. Being from the USA the ac line voltage input is specified at 110V 60Hz with a maximum power input of 500W. In the UK this line voltage requires a step-down transformer, a suitable auto-adjusting 1kW transformer was found to be adequate for long-term operation. This post presents detailed pictures, measurements, and an analysis of the HGF as a spark gap generator, and a comparison to the replica diathermy already built and reported in the Spark Gap Generator Part 1 and Part 2.

Figures 1 below show the front-panel of the unit, and in detail, the construction and configuration of the various inner components of the HGF:

The actual circuit diagram for the HGF, which was obtained by opening and measuring the unit, is shown in figure 2 below, or click here to view the high-resolution version. Click on the following link to view the original circuit diagram which appears the closest match to this unit model, and comes from the original Model G2.

The line input is connected through a switch, (also with outputs for a foot switch), to the input of a multi-tap choke. The choke is designed to restrict the current, based on the output tap setting, through the primary of the high voltage transformer, and hence control the output voltage of the transformer and the strength of the spark discharge. The model G has 5 power settings which control the input power measured in the range 75 – 440W @ 120V, (line output of the auto-adjust step-down transformer). The choke is constructed from multiple layers of windings using AWG 18 solid magnet wire, wound over thin cardboard layer separators, and coated in a Shellac type resin to hold all the windings together. The core of the choke is made from many thin mild steel laminations and mounted each end to the front panel by wooden spacer blocks.

The output of the tap selector from the choke is connected to the primary of the high voltage transformer. The primary is mounted on thicker mild steel laminated core that is rectangular in shape, with the primary at one end, and the two secondary coils at the other. The primary is wound again like the choke with multiple layers of, cardboard separated, AWG 16 solid magnet wire, held in place with the same resin, and then externally coated with paper tape. The two secondary coils are again made from many cardboard separated layers, with ~25 windings per layer of AWG 22 solid magnet wire, and structurally held together by thin wooden layers compressed by insulating fibre threaded rod, and plastic threaded round nuts.

The combination of the primary and secondary coils on the laminated steel core forms an early transformer arrangement. The efficiency of this type of basic transformer is not as high as would be expected for a typical modern design, constructed with modern materials. Increased losses in the laminated core and materials lead to more power dissipation and heating during running, although the size and bulk of the construction results in a robust transformer, that can be run all day long at maximum power without overheating. This is further evidenced that the transformer is still functioning correctly, and is not far short of 100 years old!

The two high voltage transformers are connected in series, and with the outer ends of the two coils connected to the outer ends of the 3 series connected spark gaps, via coiled cotton-clad stranded wire. The spark gaps appear to be a tungsten tipped centre electrode, clad with a machined copper heat sink fin. The other end of the centre electrode is very fine threaded, and with a bakelite adjustment handle at one end to fine adjust the gap spacing of each electrode pair. The outer ends of the 3 series connected spark gaps are connected both to the output of the secondary coils, and the inputs to the series capacitors, in the first stage of a typical Tesla[1] “hairpin” arrangement. Further detail regarding the Tesla “hairpin” circuit and its history is given in a summary presented by Kraakman[2].

The two series tank capacitors, ~15nF each, are constructed from alternating thin copper conducting sheets with thin insulating mica sheets. These are sandwiched together between two cotton webbed sheets, insulating them from the outer wooden blocks that compress the capacitor sheets together. The wooden end blocks are compressed together by insulating fibre threaded rod, and plastic threaded round nuts. In fig. 1.6 it is important to note that on the left hand tank capacitor there is another smaller capacitor formed on the outside of the wooden block. This is the floating ground connection capacitor important for diathermy use when connected to a patient. This capacitor prevents the final outputs of the diathermy unit to accumulate to very high potentials relative to ground, which would present a considerable discharge danger to the patient from their body to ground. It is somewhat disconcerting to imagine that the safety of the patient, when connected to this type of high tension generator, was only really ensured by the two series tank capacitors and the floating ground connection capacitor. Failure of any one of these three capacitors would effectively connect the patient to around 6kV @ 100mA at the input line frequency of 50/60Hz!

The two tank capacitors are connected to each end of the primary coil, which is 6 turns of 3/16” copper tube, and then extended by an 11 turn  AWG 12 solid copper wire Oudin coil extension. The low, medium, and high output taps are then derived from the primary and Oudin extension as shown in the schematic diagram of figure 2. The primary and Oudin extension are wound around a 31/2” primary former which appears to be a resin/mica composite material. The primary turns are separated from each other by a cotton woven thread which matches the diameter of the copper tube and wire. Connection between the spark gaps and the tank capacitors, the primary coil, and the output taps, are via 3/16” x 1/16” solid copper flat bar.

The secondary coil is a 90 turn 20 AWG cotton coated magnet wire on a 25/8” cardboard former, and the turns retained in place by the same Shellac type resin. The secondary coil is retained in the centre of the primary former by two wooden end caps, themselves compressed together by another insulating fibre threaded rod, with plastic threaded round nuts. The inner end of the secondary is connected to the common connection as shown in the schematic, and the other outer “hot” end of the secondary is fed through the centre of a bakelite stand-off/insulator through the front-panel and to the high tension Tesla terminal, (ball connector with 4mm socket hole). It is important to note that the secondary coil is orientated within the primary former at the opposite end to the Oudin coil extension, which prevents the very high tensions at the top-end of both the primary and secondary from breaking through the primary former forming a spark discharge path between primary and secondary coils.

The primary output taps are connected through a manually positioned thermo-ammeter shunt that shows rf output current on two ranges 0-1A, and 0-4A. The manual shunt can be positioned to remove the meter from the circuit, or to connect it in high and low range positions. The low range is protected by a 1A slow-blow fuse that shunts a 22Ω 50W wire-wound power resistor. When the fuse blows the shunt across the resistor is removed and the meter is protected by the series power resistor. The overall low-side output of the manual meter/shunt circuit, is connected to the indifferent (ground) terminal on the front-side of the diathermy main panel. The measured component parameters combined with their important physical attributes are shown on the schematic in figure 2.

Figures 3 below show the small signal impedance measurements for Z11 up to 10Mc/s at the output of the spark gaps for both the original HGF and the diathermy replica (DR) unit:

To view the large images in a new window whilst reading the explanations click on the figure numbers below, and for a more detailed explanation of the mathematical symbols used in the analysis of the results click here.

Fig 3.1. Shows the fundamental resonant frequency Fat M1 of the primary circuit, and the fundamental for the Tesla secondary coil FS at M2, and its second harmonic FS2 at M4. FP at 1120kc/s is the series resonant frequency formed by the combination of the two series tank capacitors CP connected together by the spark discharge, in series with the primary coil inductance LP. LPCP forms a series resonant circuit where the reactance of Land CP cancel each other out at resonance, leaving only the series resistance of the primary RP at M1, which in this case is 0.49Ω.

M2 shows the fundamental resonant frequency of the secondary FS = 2900kc/s, and M3 the frequency at which a 180° phase change takes place FØ180 = 3180kc/s. As is normal for a secondary coil where there is considerable distributed resistance across the coil end, points FS and FØ180 do not occur at the same frequency, and the parallel resonance formed between LS and the distributed capacitance CS set the fundamental resonance of the coil at M2. When electrical energy is coupled to the secondary from the primary the coil will resonate at the frequency indicated at M2. It is also to be noted that the Q of the resonance at M2 is considerably lower than expected, showing that losses in the secondary coil, materials, and mountings are considerable, where rf energy is being both dissipated in RS, and leaking out of the circuit formed by LSCS through parasitics to the surrounding medium. The low Q of the secondary considerably impacts the energy stored in the system, and will reduce considerably the rf oscillating currents in the secondary, which can be seen in figures 3.

The second harmonic of the secondary FS2 occurs at M4 and M5. If the primary is tuned closer to M4 then the secondary coil will resonate at FS2 = 8140kc/s which represents the second odd harmonic of the secondary wire length, 3λ/4. The parallel resonance at M4 is noted also to be very low Q, and similar in this case to the fundamental. The low Q of the secondary can most likely be attributed to, firstly, the cardboard former of the secondary coil, which over considerable time will have absorbed moisture, and presents a considerable leakage or parasitic resistance to the windings of the secondary coil. Secondly, the windings in themselves are only cotton-clad un-insulated (bare) magnet wire, which also presents a significant leakage path to a moisture impregnated cardboard former. Thirdly, the cardboard former is mounted to the primary via wooden end boards which themselves can absorb moisture, and when combined with moisture in cardboard former of the secondary could also form a significant leakage between the windings of the primary (bare copper) and the secondary coils.  It is conjectured that the Q of the secondary when the HGF was new, or much younger, would have been better than now measured, but still due to the nature of the materials used, would still present a much lower Q than that which can be obtained by using plastic formers, and with magnet wire either PTFE coated, or high temperature varnish coated.

Fig 3.2. For comparison the same small signal impedance measurement is shown from the spark gap generator diathermy replica (DR). Due to the slightly different geometric sizes, of the readily available materials used, for the primary and secondary, and the modern equivalent of the tank capacitors, the key resonant frequencies of the primary and secondary are at slightly different points. However the general characteristics of the frequency markers remains very similar on the horizontal scale.  The series resistance of the primary circuit at resonance for the DR is less than half that for the HGF, showing that larger primary currents can be generated in the DR providing stronger output currents in the low, medium, and high primary taps.

The big difference between the HGF and DR is in the Q of the secondary coil, which is very much larger, and well-defined, in the case of the DR. This shows the difference largely between the types of materials used to construct the secondary, which in the case of the DR is a plastic polypropylene former, with PTFE coated Kynar secondary windings, insulated from the plastic polypropylene former of the primary, by nylon 66 connecting bolts. All these plastic insulating mediums do not suffer with moisture absorption over time, do not degrade significantly over normal time spans, and present a very high impedance between the primary and secondary coils, which reduces any leakage currents in the secondary to very low values. Hence the Q of the secondary circuit is very sharp and well-defined.

Figures 4 below show the large signal time domain waveforms of the HGF as measured from the indicated output taps, and illustrate the different stages of the spark discharge burst both in the primary and secondary coils of the generator. The HGF was being run at an input power of ~ 300W, (monitored using a Yokogawa WT200), which was kept constant throughout the measurement. Output waveforms were measured using a Pintek DP-50 high voltage differential probe, (max. 6.5kV up to 50Mc/s), which was connected to a HP 54542C oscilloscope to observe and record the output waveforms.

To view the large images in a new window whilst reading the explanations click on the figure numbers below. For a detailed description of the different sections of the spark discharge refer to the analysis of figures 3 and 4 in Part 2 of the Spark Gap Generator post.

Fig 4.1. Shows the burst waveform measured at the low output tap of the HGF. The vertical amplitude scale is 1kV/div, and the horizontal timebase is 5µs/div. This figure shows clearly the  second section, a ring-down of specific frequency based on an exponentially decaying oscillation in the primary coil of the HGF. The third section, a ring-down of another specific frequency on an exponentially decaying oscillation in the secondary coil of the HGF is too small to be observed on this vertical scale. The maximum amplitude of the burst is ~ 5kVpk-pk, and lasts for about 7.5µs before decaying to less than 1% of its initial amplitude.

Fig 4.2. Here the horizontal timebase has been reduced to 500ns/div and the waveform buffer delay adjusted so that section two dominated with the primary oscillations fills the entire trace. The monitored average frequency of trace 1 can be seen to be 1077kc/s which is FP, the fundamental resonant frequency of the primary circuit, which corresponds well to that measured in the small signal impedance measurements at M1, of 1120kc/s.

Fig 4.3. Here the vertical amplitude and horizontal timebase have been reduced from fig. 4.1, in order to show the very small third section of discharge burst, that occurs from the secondary ring-down reflected into the primary circuit. The secondary ring-down is barely 100Vpk-pk and reflects the very low Q of the secondary fundamental resonant frequency FS, as shown in fig. 3.1. Due to the low Q from the high leakage through the materials used, the third section ring-down has a low amplitude and decays away very quickly, only lasting in this case ~ 4µs after the primary ring-down in section 2.

Fig 4.4. Here the third section has been magnified and the monitored average frequency of trace 1 can be seen to be 2929kc/s, which again corresponds well to that measured in the small signal impedance measurements at M2, of 2900kc/s.

Fig 4.5. The low tap burst, again on the original scales, for amplitude comparison with the following two figures.

Fig 4.6. Shows the burst waveform measured at the medium output tap of the HGF, and on the same scale as before. The inital burst amplitude has increased to ~ 8kVpk-pk.

Fig 4.7. Shows the burst waveform measured at the high Oudin output tap of the HGF, and on the same scale as before. The inital burst amplitude has increased to almost 12kVpk-pk.

A comparison of some of the key characteristics of the HGF and the replica diathermy already built and reported in the Spark Gap Generator Part 1 and Part 2, are shown in the following table:

H.G. Fischer Model GP Diathermy:

Primary Coil: 6 turns 3/16″ copper tube

Primary Former (OD): 31/2“, 89.6mm

Oudin extended coil: 11 turns 12 AWG solid (2mm) magnet wire

Seconday Turns: 90 turns, 20 AWG cotton-clad magnet wire

Secondary Former (OD): 25/8“, 65.7mm

From figures 3. Z11:

FP @ M1: 1120kc/s

RS @ M1: 0.49Ω

FS @ M2: 2900kc/s

FS2 @ M4: 8140kc/s

From figures 4 (large signal time domain, LSTD @ 300W):

FP: 1077kc/s

FS: 2929kc/s

VL (pk-pk): 5kV

VM (pk-pk): 8kV

VH (pk-pk): 12kV

Diathermy Replica (SGG Parts 1 and 2):

Primary Coil: 6 turns 3/16″ annealed copper tube

Primary Former (OD): 90mm (standard UK polypropylene pipe)

Oudin extended coil: 11 turns 14 AWG solid (1.6mm) magnet wire

Secondary Coil: 98 turns, 19/32 20 AWG Kynar wire (PTFE coated)

Secondary Former (OD): 63mm (standard UK polypropylene pipe)

From figures 2. Z11 (Spark Gap Generator – Part 2):

FP @ M1: 950kc/s

RS @ M1: 0.22Ω

FS @ M2: 3180kc/s

FS2 @ M4: 8390kc/s

From figures 3 LSTD @ 300W (Spark Gap Generator – Part 2):

FP: 895kc/s

FS: 3214kc/s

VL (pk-pk): 4kV

VM (pk-pk): 8kV

VH (pk-pk): 11kV

In summary, the original H.G. Fischer medical diathermy unit explored in this post is a robust, self-contained, and easy to use generator suitable for some preliminary experiments and replications in the field of Tesla and electricity research, where high tension oscillating currents are required, e.g. with a TMT experiment. The HGF’s lower overall performance, in comparison to the replica diathermy unit, largely results from age related wear and tear, component degrading, and generally lower quality, or less suitable, materials for high voltage applications. Having said this, the HGF is almost 100 years old, and still working more than adequately as a high voltage generator, which is in itself an impressive accomplishment from another time. Without access to an original HGF, it has been shown in other posts, that a good high performance generator with very similar yet improved characteristics, can be constructed using readily available materials at a very affordable cost.

Click here to continue to part 1 of the spark gap generator where the diathermy replica is designed and constructed.


1. Tesla, N., Experiments with alternate currents of very high frequency and their application to methods of artificial illumination,  American Institute of Electrical Engineers, Columbia College, N.Y., May 20, 1891.

2. Kraakman, N., A Brief History of the Tesla Hairpin Circuit, December 7, 2017, Waveguide

 

Spark Gap Generator – Part1

The complete spark gap generator (SGG), including the diathermy replica (DR), and the MMC capacitor bank unit, is shown in Figures 1 below, and mounted on top of, and connected to the high voltage supply:

In the early days of my research, and before we built the spark gap generator, it was unclear to me which parts of the electrical system were most directly responsible for generating unusual electrical phenomena, whether it be the generator or high voltage source, the types and arrangements of the various coils, or a combination of these elements setup and arranged in a specific manner, tuned in a specific way, and operated in a specific method. Far more consideration is normally given to the experimental components (e.g. coils), their construction, dimensions, materials, and the results that they yield, and much less on the generators that produce the high voltages and currents that are used to power the experimental apparatus.

Over time this appears to have led to an “air of mystery” surrounding the generators that are used in these types of experiments, quite besides a good generator is a complex and involved process to design and build, and can take much more time than any other system component to “get right”. Certainly when I started out by watching the experimental work of Dollard et al.[1,2], I was very much left with the impression that many of the unusual results obtained were mostly a product of the special generator and components used, and the experimental coils allowed these effects to be transformed, observed, and experimented with.

I have not been alone in these impressions, as I have received very similar comments from others in the field that have not actually built a working experimental system for themselves, but rather still feel that “air of mystery” that surrounds the generator and specialised components and materials in the systems construction. Only by building or contributing to a working experimental system, (including the generator), is it possible to really dispel this “air of mystery”, as it becomes possible to understand and characterise how the generator is producing the types of voltages and currents specific to its type.

In addition to this, many of the components referred to in important works such as Dollard et al.[1,2,3], may well have been more readily available in the 80s and early 90s, but are now quite scarce, and often command high prices for working items, or “new old stock” components. For example a 1920s H.G.Fischer diathermy machine used as the primary generator in the experiments of [1] are very rare, and when they very occasionally are available, they are expensive. Without understanding what is inside a generator such as this, and how it is working, it is very difficult to know how to build a comparable generator, or whether one will be able to gain the same types of unusual electrical phenomena demonstrated in works such as [1,3].

This was certainly the place I found myself in the early days where I wanted to begin by reproducing and confirming for myself the unusual measurements and results obtained by others, before using this as an established foundation to advance further in exploring my own ideas and insights regarding electricity, and the displacement and transference of electric power. When I started out there were no good examples of a working diathermy machine currently available, (especially in the UK), so I decided to design and build a diathermy replica for myself, and using readily available materials and components. If this generator could be used to explore unusual electrical phenomena then it would certainly for me increase my understanding enormously of how such generators are designed and constructed, whilst also dispelling the “air of mystery”, and providing a generator design that could also easily be used by others in the field. Later I did finally acquire an original 1920s H.G.Fischer diathermy machine  and have been able to make a characterisation and comparison between the replica and the original.

The posts reporting the Spark Gap Generator – Parts 1 and 2 are the result of building a working high power diathermy replica, which is now routinely used in my daily experiments, and has contributed significantly in bringing me to a core understanding, that all parts of the electrical system play a specific role in the generation of unusual electrical phenomena. In the exploration into the displacement and transference of electric power each part of the system apparatus must first be measured and characterised carefully to establish a well tuned and balanced overall system, and where the electric and magnetic fields of induction are balanced and in a state of dynamic equilibrium.

From this point it is possible to experiment directly into the properties of electricity, and come to an understanding that the unusual phenomena observed are a product of the inner workings of electricity itself, where the generator and experimental apparatus are necessary to set up the conditions and boundaries required to explore these inner properties.  The properties of electricity are there to be revealed rather than “generated”, which also assists greatly in dispelling the “air of mystery” that the generator is the source of the unusual phenomena, but rather, the instrument that provides the necessary tension to trigger the imbalance of the electric and magnetic fields of induction, and hence to observe electrical phenomena within the system. The differences here are subtle but hugely important to the overall understanding of electricity and particularly displacement of electric power.

The circuit diagram for the SGG and peripherals is shown in Figure 2 below, or click here to view the high-resolution version.

In the early days in order to replicate the experiments and results of Dollard et al.[1] , it was considered that the diathermy replica (DR) should be as close as possible to the original, which of course posed a challenge when there was no original from which to take measurements, dimensions, construction methods etc. To overcome this a range of references from the internet were studied, along with available circuit diagrams. The most useful references proved to be a combination of material from [3,4,5], and allowed key dimensions and some circuit component values to be extracted from the images and videos.

Click on the following links to view circuit diagrams for various original H.G. Fischer (HGF) diathermy machines, the Model G2, the Model H, the Model CDC, and the Model A[4].

In any Tesla or resonant coil design it is first necessary to define the desired properties of the secondary coil. With this defined the primary and other components of the system can be designed around the secondary properties. In the case of the DR, and in the absence of any good measurement data, e.g. the resonant frequency of the HGF secondary, the DR secondary was designed according to the known and available dimensions of the HGF secondary, the wire size and type, and the number of turns, (mostly gained from [5]). The dimensions and wire were then adjusted for readily available materials and then key parameters extracted using the software Tccad 2.0[6]. The DR secondary coil properties were adjusted primarily to match F0 the primary resonant frequency of the secondary coil, while keeping the physical dimensions as close as possible,  whilst using readily available material types and sizes. The original HGF and the designed and adjusted DR properties are shown in the following table:

Original H.G. Fischer Model G Diathermy:

From reference pictures and video[5]:

Turns: 90

Wire: Solid copper 20 AWG cotton-clad, total diameter 1.0mm

Secondary Former (OD): 25/8“, 65.7mm

 

Secondary Coil length: 92mm

Primary Former (OD): 31/2“, 89.6mm

Primary Coil: 6 turns 3/16” copper tube

Oudin extended coil: 11 turns 12 AWG solid (2mm) magnet wire

From Tccad 2.0:

L = 340uH

C = 2.87pF

F0 = 5066 kc/s

Fλ/4 = 3978 kc/s

Additional top load Cλ/4 = 1.78pF

Wire length (WL) = 18.9m

Diathermy Replica:

Adjusted for available materials to match F0:

Turns: 98

Wire: 19/32 20 AWG Kynar (PTFE coated), total diameter 1.2mm

Secondary Former (OD): 63mm (standard UK polypropylene pipe)

Secondary Former OD + PTFETAPE + PTFEWIRE = 63.40mm

Secondary Coil length: 120mm (the PTFE wire  coating adds additional length)

Primary Former (OD): 90mm (standard UK polypropylene pipe)

Primary Coil: 6 turns 3/16″ annealed copper tube

Oudin extended coil: 11 turns 14 AWG solid (1.6mm) magnet wire

From Tccad 2.0:

L = 350uH

C = 2.84pF

F0 = 5065 kc/s

Fλ/4 = 3806 kc/s

Additional top load Cλ/4 = 2.03pF

Wire length (WL) = 19.4m

Figures 3 below show how the Tesla/Oudin unit was constructed, and the types of materials used in a simple open structure that can be easily adjusted and modified according to the experimental requirements. Components are mounted on a mdf wooden base, and conductors insulated from the mdf using PTFE and Nylon 66 mounts, bolts, and nuts.

The formers of the primary and secondary coils were first coated with PTFE tape to improve the thermal barrier between the coil and the polypropylene former material when running at high output powers. In addition a small low voltage fan was located under the primary coil to help keep the primary cool at sustained high power outputs. The Tesla EHT output and the HT output taps are all mounted on PTFE insulators both for electrical isolation, but also for good resistance to melting and burning which can occur when drawing discharges from these terminals. Nylon 66 can also be used here, but has lower thermal resistance to discharges, but has the benefit of being a much cheaper material than PTFE.

It is important to note that the secondary coil is located in the primary at the opposite end to the Oudin extended coil. In an early version of the DR the secondary was incorrectly located at the same end as the Oudin extension, which because of the increased tension in both coils easily causes breakdown between the two coils, and led to burn-out of the first secondary coil. The tank capacitor banks are made from 3 series connected Cornell Dubiler 941C03 series 3kV polypropylene film capacitors each of value 47nF, which combined gives 15.7n 9kV for each bank. These tank capacitors have been proven to be long-lasting, and robust, and have never been changed,  even with input powers in excess of 1.5kW and even up to 2.2kW for short bursts of power.

The tank capacitors are force air-cooled, and mounted on insulated conductors which allow for easy connection and adaption to the circuit under test. The capacitance of the tank was initially higher at around 47n to match more closely the HGF schematic of the Model A, but was reduced to its current value after the real HGF was acquired and measured. At 47n the available output power was quite a bit less than with 15.7n per tank, as the primary resonance is pushed lower and further away from that of the secondary, hence reducing the primary currents, and hence the power coupled from the primary to the secondary.

Figures 4 below show how the static spark gap (SSG) unit was constructed.

The SSG unit was at the time the most difficult unit to build when only a large pillar drill, large vice, and bench sander were available in the workshop for mechanical construction. Each electrode of the spark gap is made from 1/4” diameter 99.9% pure tungsten rod 1″ long, (not to be confused with tungsten carbide rods), which were pressed using the vice, into the centre of a drilled A2 stainless steel fine pitch bolt. The bolt had a 6.2mm hole centre drilled, by mounting the bolt in the drill chuck and clamping the drill stationary on the stage, and opposite to how a drill press is normally used. This arrangement made a very rudimentary “lathe” and made it easier to drill a centralised 1.5″ hole down the centre of the bolt. The tungsten electrode was then pressed into the bolt leaving 5mm externally for the spark electrode. Alternating stainless steel washers (large and small) were then threaded onto the bolt and finally tightened with a thin stainless steel nut to form the cooling fins of the electrode body.

Each pair of electrodes were then mounted in threaded aluminium blocks, locked in place with a nut on one side, and with a threaded bakelite handle on the other, to allow adjustment of the spark gap space by winding the bolt in or out of the aluminium block. The aluminium blocks were arranged and mounted to form a series connection of all four spark gaps, and also allowing for tapping from any of the 4 stages for experiments using a single gap, all the way up to 4 series gaps, or 4 parallel gaps with shorting shunts. The adjustable electrode was tensioned in the alumium thread by a small compressed nylon rod, ( from an M3 nylon bolt), which was inserted in a vertical hole drilled above the thread, and then tensioned using a screw locked in the correct place by a nut.

The aluminium spark gap blocks are mounted onto PTFE insulators and then mounted to a wooden base where each gap is suspended above force cooling provided by a pair of low voltage plastic fans mounted into the base of the unit. Overall the SSG unit is robustly constructed and can withstand very large powers in constant use. Tuning of the gaps for optimal running can be made carefully during operation via the four insulated adjustment handles, and is demonstrated in the operation video in Part 2.

Figures 5 below show the MMC tank capacitor bank (TCB) which was used to remove one of the tuned primary stages, and hence increase the efficiency and optimisation of the generator driving, for example, the tuned primary of a TMT (Tesla magnifying transformer).

The SGG used with the DR as a generator is most commonly connected with one of the L, M, or H outputs to the primary stage of a tuned TMT, of which the primary stage would typically consist of a coil with a parallel tuning capacitor. In this arrangement the DR, which in itself is already a tuned primary stage, is now connected to another tuned primary stage. Whilst this was useful for preliminary experiments in confirming the key experiments and results of Dollard et al.[1,2] and keeping the generator as close as possible to an original HGF, it has been shown to be more efficient to eliminate the double tuned primary stage by removing the DR and placing the TCB in its place.

The TCB is simply a standalone capacitor unit exactly the same design as used in the input to the DR. Here the TCB is shown with two pcb MMC banks, but the Cornell Dubilier 941C03 series banks could equally as well be mounted in the same way. Figures 1.6 and 1.7 show how the TCB is connected directly to the SSG, and then in turn the output of the TCB is connected directly to the input of, for example, the tuned primary of a TMT, or some other experiment or load. This creates a resonant drive circuit with the TMT coil in series resonance with the TCB capacitors, and is exactly the same arrangement as internally for the DR, and the HGF. Use of the TCB in replacement for the DR has allowed for an easier and more accurate resonant match between the SSG unit and the TMT load, and with greater power transfer between the two units. Impedance measurements are also simplified by removing one resonant circuit from the generator chain. A very wide range of capacitors can be mounted on the TCB, and for higher power experiments up to 2.5kW output power, force cooling is available via the low voltage fan in the base of the unit.

Overall the spark gap generator whether used with the DR unit or the TCB unit at the output, has proven to be a robust and reliable generator for a wide range of experiments. It has enabled the replication of the key experiments as presented by Dollard et al.[1,2,3,4], as well as forming a flexible and powerful tuned static spark generator for my own experiments in the displacement and transference of electric power, as well as telluric transmission experiments.

Click here to continue to part 2 of the spark gap generator where the operating characteristics are measured both in frequency and time, as well as a short video to show its general operation and running.


1. Dollard, E. & Lindemann, P. & Brown, T., Tesla’s Longitudinal Electricity, Borderland Sciences Video, 1988.

2. Dollard, E. & Brown, T., Transverse and Longitudinal Electric Waves, Borderland Sciences Video, 1988.

3. Dollard, E. & Mackay, M., Tesla Radiant Energy Experiment, Bedini-Lindemann Conference, June 29-30, 2013.

4. Mackay, M. & Dollard, E., Tesla’s Radiant Matter Replication, 2013, Gestalt Reality

5. Sergey Z., Fischer Diathermy Narrating and Exploring a 1920’s Tesla Coil, May 2014, Youtube

6. Chapman R., Tccad 2.0 for Windows, 2000.

 

Spark Gap Generator Measurements – Part2

Part 1 of the spark gap generator covered the major components of the system, along with the design steps taken to build a diathermy replica unit (DR). In this part measurements are carried out both in the frequency and time domains, to further understand the operating characteristics, and how best to match the output of the generator to the experimental load. In this part there is also consideration as to how the generator transforms the incoming mains supply to an output suitable for experiments in the displacement and transference of electric power.

The primary purpose of any generator within such an experimental system, arranged to investigate the inner properties and workings of electricity, is to provide the necessary tension to the experiment, in order to change the balance of the electric and magnetic fields of induction within the local region of the experimental system. It is considered that changing the local balance of these fields in turn couples to deeper properties within the energetic dynamics and wheel-work of nature, which according to the purpose or the load of the system generates a response into the local experimental system. In so doing the form of the electrical input to the generator is transformed to another more  suitable electrical output under tension. In other words the energetic balance of the system is based on an inter-dependence between the local source, (in this case the generator), and the “need” or purpose generated in the system, (in this case the load). The inter-action between source and load defines the local electrical characteristics of the system under experimentation.

In the case of the spark gap generator tension is established by considerably raising the potential (voltage) of the output, whilst simultaneously transforming incoming alternating currents (ac), to oscillating currents (oc) in both the primary and secondary coils in the DR. In addition, and most importantly for displacement, there is a brief moment before the initiation of the discharge of the spark where the impedance of the space in the gap is low, but no transient discharge has yet started. At this point it is conjectured that displacement occurs, and an impulse current is drawn into the system for a very brief moment before the spark discharge is established.

After this moment of displacement, current starts to flow from the tank circuit through the spark gaps, dissipating the stored energy in the circuit through the normal process of transference, and in so doing generating oscillating currents in the resonant circuits of the primary and secondary. It is conjectured that exploration of these transient impulse currents may indicate a mechanism for additional energy to be injected into the system, and is part of the larger displacement principle being investigated as an inner working of electricity, and originating from the undifferentiated coherent action of the electric and magnetic fields of induction to re-balance the dynamics of the local system.

Figures 2 below show the small signal impedance measurements for Z11 up to 10Mc/s at the output of the spark gaps, (with the HV unit disconnected), and then with progressive change of tank capacitance to show the change in tuning, and the optimum match between the primary and secondary of the diathermy replica (DR) unit:

To view the large images in a new window whilst reading the explanations click on the figure numbers below, and for a more detailed explanation of the mathematical symbols used in the analysis of the results click here. For further detail in the analysis and consideration of Z11 typical for a Tesla coil based system click here.

Fig 2.1. Shows the resonant frequencies of the both the primary and the secondary coils in the DR. M1 (marker 1) is the fundamental resonant frequency FP of the primary, showing the 180° phase change that takes place at the resonant frequency, and the minimum impedance point of a series resonance where |Z| has no reactive components and only reflects the electrical resistance of the primary coil. FP at 950kc/s is a result of the series combination of the primary coil inductance and resistance LP and RP, and the combined two series banks of tank capacitors CP, and any stray L and C that result from the inter-connecting wires and boundaries to the surrounding medium. RP ~ 0.22Ω is low and indicates a good primary coil size and material, which will enable larger discharge currents to flow, facilitating stronger oscillations to be coupled to the secondary, and an improved power transfer between the primary and secondary coils.

M2 shows the fundamental resonant frequency of the secondary FS = 3180kc/s, and M3 the frequency at which a 180° phase change takes place FØ180 = 3820kc/s. As is normal for a secondary coil where there is considerable distributed resistance across the coil end points FS and FØ180 do not occur at the same frequency, and the parallel resonance formed between LS and the distributed capacitance CS set the fundamental resonance of the coil at M2. When electrical energy is coupled to the secondary from the primary the coil will resonate at the frequency indicated at M2.

Where required FS can be made to more closely match FØ180 by adding additional loading capacitance to the open end (top-load) of the secondary coil. This is a very common practice for large discharge Tesla coils, (designed for powerful streamers), where metal toroids are added as a top-load and add additional loading capacitance bringing FS much closer to FØ180. This also reduces the Q of the Tesla coil and hence is not desirable for experimental coils designed to explore the inner workings of electricity.

It should be noted that the fundamental resonant frequencies of the primary FP and secondary FS do not correspond at the same frequency, as would normally be expected and tuned for a spark gap driven Tesla coil arrangement. Normally to gain maximum power transfer between the two coils their resonant frequencies will be arranged to be the same through tuning of the primary, (Lp or Cp dependent on the type of coil, how it is constructed, and with what materials). This means that in the DR case less power than optimum is coupled to the secondary, and hence the strength of any discharges from the output of the EHT terminal are reduced. Since the DR is based on the original HGF specifications it is conjectured that this may have been desirable for medical diathermy applications to restrict the strength of the EHT discharges by deliberately mis-matching the resonance of the two coils.

The second harmonic of the secondary FS2 occurs at M4 and M5. If the primary is tuned closer to M4 then the secondary coil will resonate at FS2 = 8390kc/s which represents the second odd harmonic of the secondary wire length, 3λ/4. The parallel resonance at M4 is noted to be quite strong, with a similar Q to the fundamental, indicating that the secondary could have a better response to impulse currents generated in the system. Impulse currents due to there very sharp, high energy, wide frequency band, excite a wide range of resonances within a typical Tesla coil system. The ability for the system to respond to such impulse currents largely depends on the overall Q of the coil’s harmonics. The series resistance at M5 becomes the limiting factor in how much power can be coupled to harmonics of the coil, and has risen considerably from M3 from 2.3Ω to 10.8Ω.

It can be noted from part 1 that the designed Fλ/4 (FØ180) was simulated for the coil dimensions, turns, and construction as 3806kc/s which is only ~ 0.4% error from that measured in the small signal Z11 analysis at 3820kc/s, (Fλ/4 occurs at M3, and is based on the λ/4 length of the coil when one end of the coil is at a low impedance, and the other at a high impedance).

Fig 2.2. Shows the dramatic effect of reducing the total tank capacitance CP down to 250pF. The marker number for the primary M1 has been kept the same despite the order of the coil resonances changing across the 10Mc/s band. M1 the series resonant frequency of the primary FP has now moved right up to 5Mc/s, which has also resulted in a reversal of M2 and M3 so the that FS is now above FØ180 at 3150kc/s. The effect of moving the primary resonance point, through the tuned primary tank CP, is to mis-match the primary and secondary resonances the other way, increase the effective series resistance of the primary coil resonance from 0.22Ω to 2.0Ω, but to leave the actual fundamental resonance frequency of the secondary FS with only a ~1% change from 3180kc/s to 3150kc/s. Increasing M1 to between the fundamental FS and the second harmonic FS2 has also had a more dramatic impact on the  frequency of the second harmonic, reducing it from 8390Kc/s to 8200kc/s, a change of ~ 2.3%.

It should be noted that the dependence of FS and FØ180 to tuning in the primary is dramatically different for the flat coil parallel tuned, and the cylindrical case series tuned. For the flat coil, parallel resonance tuned, FØ180 remains more constant with changes in CPP, and is almost exclusively effected only by the secondary wire length, whereas FS, and its harmonics FSN, vary very widely based on changes in CPP. In the cylindrical coil, series resonance tuned, the dependence reverses and FØ180 varies very widely with changed in CPS , whilst FS, and its harmonics FSN, remain more constant with changes in CPS. This emphasises the need for the correct choice in the type of secondary coil used for any specific experiment (e.g. flat, cylinder, equal ratio etc.), and also the correct choice of primary tuning mechanism, (parallel or series). The characteristics and differences, and hence the choice for specific types of experiments, for each of these different coil configurations will be considered and reported in more detail in subsequent posts on the cylindrical coil.

Fig 2.3. Here CP is now increased to 500pF and M1 starts to move downwards again towards the secondary FS. In this case FP is approaching the point of optimum match where the primary and secondary are equally split between the centre point. With CP = 500pF the match is still a little high where the primary is resonating at a frequency above the secondary.

Fig 2.4. Here CP is now increased to 750pF and M1 and M2 are now equidistant either side of FS at M3. Once again FS has not really changed significantly and is still at 3150kc/s. This point of match is in principle the most optimum match between the primary and secondary coils, where maximum power can be transferred between the two coils.

In practise and for maximum streamers it is usually preferred to operate this form of cylindrical Tesla coil, where FP is slightly below FS, due to FS falling when a discharge (streamer) occurs. The discharge causes a change in the impedance of the secondary coil reducing its resonant frequency FS, bringing FS during discharge to an optimum match with the primary, allowing maximum power transfer from the tank through to the secondary discharge.

In experiments to explore the displacement and transference of electric power, where it is preferable not to produce discharge streamers (dissipating the energy of the system through transference), the optimum match where FS = FP is the preferred condition. This is where the Q of the system is maximum, and the continuity between the electric and magnetic fields of induction between primary and secondary are optimum, which in turn ensures the maximum dynamic stability, and best departure point from a system in equilibrium.

Fig 2.5. Here CP is now increased to 1000pF, Fis slightly below FS, (observed in the larger gap between M1 and M2, than M2 and M3), which is around the best empirical match for a Tesla coil designed for maximum discharge as discussed in the previous section.

Fig 2.6. Increasing CP to 2000pF starts to move Fmore rapidly away from FS, the match between the primary and the secondary is reducing, and hence the coupled energy is also reducing.

Fig 2.7. At CP = 5000pF FP is now approaching the DR design of Fig 2.1, FP = 1000Kc/s, and FS remains mainly constant at 3140kc/s, only having changed ~ 0.3% as CP changes in the range 250pF – 5000pF.

Fig 2.8. At CP = 5500pF FP is now very similar to the DR design of Fig 2.1, however FS has not yet increased slightly to match the 3180kc/s in Fig 2.1. CP is somewhat different to the expected ~ 7200pF of the two Cornell Dubiller tank capacitor banks which in combination is 6 capacitors of 47nF in series.

Fig 2.9. Here CP has been increased to 6100pF, where FP matches to the large signal primary resonant frequency observed during the time domain experiments shown below in Fig 3.3 at 895kc/s. FS which is now 3190kc/s has finally moved slightly away from the previously stable 3150kc/s, but notably is now closer to the DR design of Fig 2.1, and also the large signal secondary resonant frequency of 3214kc/s shown in Fig 3.6 below.

Overall the small signal Z11 analysis of the spark gap generator reveals a wealth of detail in understanding how this generator is characterised in the frequency domain, and how best to match the primary tank capacitance to obtain different operating points according to the purpose of the experimental system.

Figures 3 below show the large signal time domain waveforms of the spark gap generator as measured from the low output tap, and illustrate the different stages of the spark discharge burst both in the primary and secondary coils of the generator. The spark gap generator was being run at an input power of 300W, (monitored using a Yokogawa WT200), which was kept constant throughout the measurement of both Figures 3 and 4. Output waveforms were measured using a Pintek DP-50 high voltage differential probe, (max. 6.5kV up to 50Mc/s), which was connected to a HP 54542C oscilloscope to observe and record the output waveforms.

To view the large images in a new window whilst reading the explanations click on the figure numbers below:

Fig 3.1. Shows the burst waveform measured at the low output tap of the DR. The vertical amplitude scale is 1kV/div, and the horizontal timebase is 5µs/div.  The oscilloscope was adjusted to acquire the burst in single-shot mode, triggering at a low to high transition of 1.75kV, where the output delay was adjusted to coincide the trigger to the start of the second full horizontal division. The burst waveform is formed of three major sections, where the first is right at the point of triggering and always involves a very sharp impulse type transition, the second a ring-down of specific frequency based on an exponentially decaying oscillation in the primary coil of the DR, and the third, a ring-down of another specific frequency on an exponentially decaying oscillation in the secondary coil of the DR.

The first section occurs right around the moment of initiation of discharge of the spark gap, and includes a very sharp impulse transition, where the amplitude of this impulse can be many times more than the nominal tension of the high voltage supply. This section  requires more detailed capture and measurement with a more sophisticated experimental setup, and will therefore be considered and reported in a subsequent post. Here it is sufficient to understand that there is an impulse like start to the spark discharge, which only lasts for a very brief moment around the initiation of the discharge, and produces very narrow and sharp amplitude spikes at the very beginning of the output burst.

The second section is established right after the spark discharge has started, and the energy stored in the two tank capacitor banks is being discharged in the primary circuit. Before the spark discharge is initiated the tank capacitors are charged by line frequency alternating current supplied by the output of the high voltage supply, where the charging circuit is formed by the high voltage transformer connected through the tank capacitors to the primary coil. When the tension across the output of the transformer has risen above the combined breakdown voltage of the spark gap unit, the spark discharge begins and the impedance across the spark gap suddenly changes from an open-circuit to almost a short-circuit.

The inputs to the primary tank capacitors are now shorted together by the spark and the tank capacitors discharge their stored energy rapidly through the primary coil. The resonant primary circuit formed by the tank capacitors in parallel with the primary coil cause the discharge to oscillate at a frequency defined by LPCP, and this oscillation lasts until the tank capacitors are completely discharged. How rapidly the capacitors discharge at the resonant frequency and the magnitude of the oscillating currents generated in the primary circuit is dependent on the series resistance presented by the primary circuit, which should ideally be as low as possible, and in the case of the DR was measured in Fig 2.1. to be ~ 0.2Ω.

The oscillating currents in the primary during the spark discharge of the tank capacitors, couple through induction to the secondary of the Tesla coil in the DR, or more clearly, a sudden change to the prior equilibrium state of the electric and magnetic fields of induction energy in the system result in energy being accumulated in the secondary coil. In the third section of the burst discharge this accumulated energy in the secondary transforms to oscillating currents at a frequency defined by the secondary resonant circuit LSCS. The secondary oscillating currents decay exponentially in the secondary coil, (assuming no streamer discharge from the secondary), according to the series resistance presented in the secondary circuit. These secondary oscillating currents again couple through imbalance in the electric and magnetic fields of induction back to the primary circuit, where they can be observed in the output waveform as the third section of the ring-down, which dominates the output when the second section oscillations have become sufficiently small.

The complete burst waveform lasts for about 20µs before decaying to less than 1% of its initial amplitude. Bursts are initiated each new cycle of the line frequency, so for UK standard line input at 50Hz to the high voltage supply, a burst is generated every 10ms, (2 per cycle), or at a frequency of 100Hz.

Fig 3.2. Here the horizontal timebase has been reduced to 2µs/div which accordingly magnifies the burst discharge showing more detail in the first, second, and third sections. In the second section and with careful observation it can be seen that the oscillation is not a pure sine wave, it is actually the oscillating currents of the primary circuit with the smaller oscillations of the secondary super-imposed over the top. The super-imposed secondary currents are not easily discernible in the second section because the amplitude of the oscillation in the primary circuit are large.

As these primary oscillations decay away, and after ~ 8µs, a phase change in the output occurs and the secondary oscillations now dominate the output with an envelope that carries the small decaying primary oscillations. In other words the overall burst waveform is a superposition of the oscillating currents in both the primary and the secondary in both sections two and three, where one or the other can be clearly observed based on the energy stored in the respective resonant circuit, and that coupled forward and backward through the inter-action  of the two coils.

Fig 3.3. Here the horizontal timebase has been further reduced to 1µs/div and the waveform buffer delay adjusted so that section two dominated with the primary oscillations fills almost the entire trace. The transition to the third section can be seen in the last two divisions of the trace. With section two the main focus of this trace the monitored average frequency of trace 1 can be seen to be 895kc/s which is FP, the fundamental resonant frequency of the primary circuit. The amplitude of the primary oscillations is almost 4kVpk-pk at the beginning of the section, and has decayed after 8µs to ~ 1kVpk-pk.

Fig 3.4. Shows the discharge burst in magnified amplitude against the original horizontal timebase rate. The amplitude has been magnified by a factor of 10 from 1kV/div to 100V/div which illustrates clearly the transition to the third section where oscillations in the secondary coil, coupled back into the primary coil, are a superposition of the both the primary and secondary oscillations, and hence the envelope of the waveform in the third section appears similar to an amplitude modulated waveform. Note: the indicated frequency on this trace is not accurate as it is calculated by averaging together sections 2 and 3, and cannot be considered to be the fundamental frequency of the secondary coil.

Fig 3.5. Here the discharge burst is magnified both in vertical amplitude and in the horizontal timebase, and illustrates more clearly the decay and envelope of the secondary oscillations.

Fig 3.6. Here the discharge burst is further magnified in the horizontal timebase and delayed into the third section of the discharge burst, which shows the monitored average frequency of trace 1 to be 3214kc/s which is FS, the fundamental resonant frequency of the secondary coil.

The large signal time domain waveforms have also revealed a wealth of detail about the operating characteristics of the spark gap generator, showing the nature and characteristics of the oscillating output waveform, and with well-defined sections that can be corresponded to the frequency domain properties measured in Figures 2.  The results have also shown impulse like characteristics in the first section of the waveform, that certainly require more investigation and more detailed measurement to clarify if they relate to, and contribute to, the conjecture of underlying displacement phenomena within electricity.

Figures 4 below show a comparison on the same vertical and horizontal scale of the low, medium, and high output taps. It can be seen that the amplitude of the output increases with each successive tap, consistent with the geometry of the primary/Oudin arrangement of the coils.

The low tap produces about 4kVpk-pk initial output, the medium tap 8kVpk-pk, and the high Oudin tap ~ 11kVpk-pk but at considerably reduced current. The medium output tap has been determined to be the best tap for driving TMT experiments, and other experimental apparatus suited to the exploration of the displacement and transference of electric power, where there is high output tension combined with stronger oscillating currents.

Summary of the generator results and conclusions so far:

1. The results and measurements for the spark gap generator correspond well between the frequency and time domain, and give a good insight into how this type of generator works, and the type of output that can be generated. The generator presented in parts 1 and 2 were initially used to confirm the experiments and results of Dollard et al.[1,2], before being applied widely to my own research into the inner workings of electricity.

2. This generator has been proven to be reliable and robust and can sustain indefinitely output powers of 1.5kW, and short bursts over 2kW with the appropriate connections and arranged loads.

3. This generator transforms the low frequency alternating currents of the line input, into high frequency oscillating current outputs, combined with considerably increasing the tension of the output.

4. Analysis, of particularly the time domain results, indicates a first section in the discharge burst that may include impulse currents and effects that are conjectured to involve displacement events. This section requires more detailed measurement and analysis, and will be reported in subsequent posts.


1. Dollard, E. & Lindemann, P. & Brown, T., Tesla’s Longitudinal Electricity, Borderland Sciences Video, 1988.

2. Mackay, M. & Dollard, E., Tesla’s Radiant Matter Replication, 2013, Gestalt Reality