Sooner or later research into the underlying nature and principles of electricity must inevitably lead to those larger philosophical and esoteric questions surrounding the origin and purpose of life, its mechanisms that constitute the wheelwork of nature, and our purpose and part to play as very small cogs in this grand design. I have in previous posts started to tentatively touch-on and develop my own current understanding of the wheelwork of nature through ideas, designs, experiments, and conjectures regarding displacement and transference of electric power. This post is the first in a sequence looking at experiments in electricity which reveal or suggest clues about this underlying wheelwork, with the associated phenomena and results, their possible origin and purpose, and how we may form a synchronicity with this wheelwork, and hence benefit from a journey that increases our knowledge and awareness of our-self and that of the great mystery or grand design. This first post in the series looks at the wheelwork of nature - fractal "fern" discharge experiment, along with observations, measurements, and interpretation ... Read post
Some of the most fascinating areas of research into the inner workings of electricity, are those that display unusual and interesting phenomena, and especially those not easily understood and explained by mainstream science and electromagnetism. The field surrounding Tesla's radiant energy and matter, the apparatus, experiments, and wealth of unusual electrical, and even non-electrical related phenomena, is a particular case to note. This first post in a sequence serves as a practical and experimental introduction to this area, along with consideration and discussion of the observed phenomena, and possible interpretations as to their origin and cause ... Read post
In this post we take a preliminary experimental look at the transference of electric power using a cylindrical coil TC and TMT, energised using a linear amplifier generator, and also the high power transfer efficiency that can be achieved in a properly matched system. The setup, tuning, and matching of the linear amplifier is covered in detail in the video experiment where a 500W incandescent lamp can be fully illuminated at power transfer efficiencies over 99% in the close mid-field region. The power is shown to be transferred to the receiver through a single wire between the transmitter and receiver coil through the longitudinal magneto-dielectric mode, and not through transverse electromagnetic radiation or through direct transformer induction. This high-efficiency, very low-loss transference of electric power is possible as the dielectric and magnetic fields of induction are contained around the single wire ... Read post
In this second post on the Tube Power Supply series I present a complete design and implementation of a high voltage (HV) unit suitable for use as a high-power plate supply, and also as a general purpose high-tension source for a wide range of experiments in electricity. I use this unit extensively in my own day-to-day research for experiments in the displacement and transference of electric power. The 5kW high voltage and plate supply is based around three heavy-duty industrial 1.8kVA microwave oven transformers which can easily be inter-connected in a range of different parallel and series configurations. The transformers can be easily combined with different output stages including a bridge rectifier, level shifter (doubler), and a high voltage discharge unit, which are all incorporated into the complete housing of the supply. The complete high voltage supply is housed in a traditional varnished wooden enclosure and is designed to fit together with the other supply components in the tube power supply series.
Note: A high voltage supply is capable of delivering voltages and currents, even at lower powers, that are instantly lethal, and that any design and operation of a high voltage unit should be undertaken with great care by a trained and experienced individual. The high-voltage supply presented in this post is intended for high-power electricity research experiments undertaken by trained and experienced operators only. The different transformer configurations combined with the different output stages make for a very versatile, robust, and adaptable high voltage and plate supply with a fully loaded output ranging from 2.1kVRMS @ ~ 2.3A all the way up to 15kVRMS @ ~ 150mA. This very wide output range currently accommodates all of the tube amplifiers, oscillators, and impulse generators that I use in my own research, including the following examples that are used in experiments presented, or yet to be presented, on this website:
1. A basic parallel connected quad 811A linear amplifier or Hartley power oscillator, using 1.2kV plate supply and producing about 1kW of sustained output power at frequencies up to ~4Mc.
2. A parallel connected dual 833C class-C Armstrong oscillator using a 4kV plate supply and producing up to 2.5kW of sustained output power at frequencies up to ~4.5Mc.
3. A single GU5B class-C Armstrong oscillator using a 4-5kV plate supply and producing up to 2kW of sustained output power up to ~3Mc, or even using a 9kV plate supply when used in pulsed-mode with a low duty cycle.
4. A dual push-pull connected 4-400A linear amplifier using 4kV plate supply and producing up to 1kW of power up to ~5Mc.
5. A dual 5C22 hydrogen thyratron pulse generator, with an anode supply up to 15kV.
Figure 1 below shows a summary table of the main setup configurations that can be arranged with the presented power supply, and the nominal outputs that can be achieved using that configuration, and with the various indicated output stages. These performance characteristics are presented as a guide to the configuration and usage of this high voltage supply, and may vary according to the type of load or generator being driven, the impedance match conditions between the supply and the generator and experiment, and also the type and condition of the transformers used in the supply build.
The following video takes a detailed look at the high voltage plate supply, its design, development, and implementation, how to configure and setup the required operation mode, the different output stages, the various safety requirements during its operation, and concluding with a demonstration of its operation during experiments in the Wheelwork of Nature series, when used with the single GU5B class-C Armstrong oscillator generator.
Figures 2 below show the high voltage and plate supply in detail both from the exterior panels and sides, through to the internal modular boards, layout, and construction.
Figures 3 below show the complete circuit diagrams for the high voltage and plate supply across three sheets. The high-resolution versions can be viewed by clicking on the following links Fig 3.1, Fig 3.2, and Fig 3.3.
Principle of Operation – General Summary
In principle the plate supply is very simple consisting of three microwave oven transformers that can be easily connected in a variety of parallel and serial configurations. Power is provided to the transformers from the line supply and via a high power SCR control unit equivalent to a powerful light dimmer control, or from an external source such as a variac or other type of power controller. The selected line supply is then fed to the three transformer power switches on the front panel. These three position switches have a centre off position, and then on position either both up and down. The on positions are arranged to swap the live and neutral connections to the transformer so changing the phase of the line supply to each transformer. The change in phase of the line supply allows transformers to be configured in different arrangements both as positive and negative output with a centre ground point. This is particularly useful in the case of the three series transformers where maximum voltage from core to primary needs to be restricted. This is covered in detail in a later section below.
Phase controlled line supply is then fed from the front panel to the transformer board primary coil circuits. The patch board allows for configuration of the connections on the secondary coil side of the transformers. The output of the patch board feeds various different modules including direct output, the HV bridge rectifier, and the HV level shifter. The selected module is finally connected to the final high tension (HT) output via a second patch board on the HT rear output panel. The HT output is then also connected to the HV discharge board, and also to the HV output monitor board. The HT output board also provides intermediary connections for tank/blocking capacitors facilitating the series and parallel connection of large HV capacitors safely and in close proximity to the power supply outputs. The wooden enclosure is so arranged to accommodate other devices in the tube power supply series, as well as open access to the main components through a large access panel in the top of the plate supply. In extreme operating and prototype conditions I often run with this access panel open (and via remote control) in order to watch for any unusual or unexpected effects.
The complete supply is housed in a varnished wooden casing, and internally arranged and assembled to be easy to repair, maintain, and modify. Module boards can be easily removed internally, and side panels fold open whilst still electrically connected for easy measurement and fault diagnosis. It should be noted that this type of power supply is designed for research prototyping and hence encounters a very wide range of different loading and matching, all the way from an open circuit condition on the output, through to heavily overloaded current conditions, and very high reflected RF and transient power conditions. These extreme operating conditions necessitate that the power supply is easy to diagnose, adjust, and repair internally, and hence it is arranged and assembled accordingly with easy access to all critical internal systems.
Power Input and Control Panel
Fig 3.2 shows the circuit diagram for the Line Supply, Power Control, Low Voltage Supplies, and Cooling. The line supply from the rear input panel provides both line supply outputs for chained connection of other modules, devices, and instruments, and internally splits into two feeds, one for the HV supply, and one of the low voltage (LV) supply. The LV supply is fused and switched with a LV indicator to show active operation. The LV line supply feeds a 15V 3A switched mode power supply which powers all the internal LV components, and also has external outputs to power other LV devices and modules in the experimental setup. Internally the 15V is stepped up to 24V by a DC-DC converter. The 24V is suitable to switch the W1W or B1B vacuum relays which operate quickly and reliably at the higher voltage. The internal cooling fans are both switched and are powered from the 15V LV supply. The fans are especially necessary during prolonged high power usage, and are positioned directly behind the HV transformers.
The HV supply is protected by a dual-pole 32A MCB, (upgradeable to 40A MCB in extreme conditions), and with a neon indicator to show active operation. The HV line supply is fed directly to a 250V 10kW SCR which is arranged for both internal and remote control via the front panel. The SCR provides progressive power control for HV transformers which is often most necessary for microwave oven transformers that have had their magnetic restriction shunts removed. The SCR voltage profile is also highly non-linear which in some experiments like Tesla’s Radiant Energy and Matter, and Displacement and Transference of Electric Power series, is most useful to reveal, accentuate, and maximise certain types of phenomena including displacement and radiant energy, and dielectric induction field charging and storage. The SCR output is fed to a line supply selection which switches either the SCR output or external line supply input to the power control front panel. The line supply selection was included to allow for quick switching to a variac for progressive linear control of a sinusoidal line supply which is most useful during experiments with phenomena that vary with supply voltage profile.
Fig 3.1 shows the circuit diagram for the Power Input Control Front Panel which consists of the following:
1. The three off and phase control transformer switches. Line supply from the selection switch on the main power panel is fed to the switches each with three positions, off and up and down, where the up and down positions switch the line supply to the respective transformer, and also swap the live and neutral from up to down to control phase control of the transformer primary. Each switch is accompanied by a neon indicator that both shows if the transformer is currently active, and the intensity that the transformer is being driven.
2. The digital power monitor is 9V battery driven in order to have independent operation from the line supply, and continues to be active even if the line supply is removed, this is important for safety in the event of a fault where the line supply is still connected to the rear panel, but has become disconnected internally due to say an SCR open circuit fault, and line supply to the transformers can still be monitored. The digital power monitor takes input directly from the line supply fed to the front panel for voltage, and for current via a current transformer in the neutral line supply return, and mounted to the inside of the power control panel. The meter has an on-off switch P-On, and is also angled upwards in the panel for easy reading. The meter provides a useful real-time summary of all operating measurements on the line supply side, including apparent voltage and current, real power indication and consumption, and total power factor.
3. The neutral line supply return also includes a 30A AC meter which is particularly good for quick monitoring of the total transformer primary current. This is useful in high current drive scenarios when changes in tuning can easily place the power supply in a very different operating condition, where very large currents are suddenly drawn from the line supply e.g. whilst tuning through the transition between the lower and upper parallel resonant modes of a Tesla coil whilst driving at moderate to high-powers > 1kW.
4. A locking switch to change from the internal SCR control potentiometer to the external remote control potentiometer. The switch is locking in order to avoid accidental switching which could yield dangerous and unexpected results if the SCR suddenly was switched to a higher power condition. The selected potentiometer connects directly back to the SCR on the main power panel and controls progressively the active portion of the line supply cycle that is fed to the transformers.
5. The remote control socket is a 10-pin connector which currently has 2 lines for the SCR remote potentiometer, and 2 lines to switch the HV discharge module on and off. The other 6 lines are not used and available for future expansion and functionality.
Transformers and Patch Board
Fig 3.1 shows the circuit diagram for the Transformers and Patch Board. The microwave oven transformers (MOT) are a heavy-duty industrial type rated to 1.8kVA with the magnetic shunts removed. A traditional MOT is a cheap high voltage transformer manufactured with the minimum weight of copper and hence cost, and designed to match the very specific impedance of a magnetron when correctly matched using the level shift capacitor. The cheap construction of the transformer usually involves welding the laminated metal core together on both sides, which whilst simple to make, results in shorting out much of the laminated core reducing it electrically to a large block of magnetic material that will easily saturate when sufficient power is applied to the primary coil. In this basic form the MOT does not easily lend itself to a progressive linear power supply at high voltage, like other types of high voltage transformers. The MOT however does benefit from being very robust and also able to supply high currents up to easily 1A at around 2kV AC.
The magnetic shunts are so arranged during manufacture of the MOT to reduce the free magnetic coupling between the primary and secondary coils, and hence limit the power transfer from primary to secondary, driving the magnetron impedance efficiently, without core saturation and hence excessive heat generation, and without pulling excessive current from the line supply. When reused as a high voltage transformer in this type of plate supply the magnetic shunts restrict significantly the maximum power output performance of the transformer, and need to be removed carefully (to avoid damaging the windings), with a drift and heavy mallet. I made up a wooden jig screwed to the bench to hold the transformers securely whilst driving out the magnetic shunts. The un-shunted MOT now benefits from no restrictive magnetic coupling, but does now need to be current limited to prevent excessive core-saturation at the top-end of the line supply input, and with higher impedance loads at the output of the generator e.g. a vacuum tube generator.
Current limiting can be achieved a variety of ways, including chokes in the primary and/or secondary coil circuits, but in this plate supply I use an SCR power controller which provides progressive power output by varying the active line supply cycle. The SCR introduces large non-linear distortion in the line supply to the transformers which is both a hindrance in some experiments and requires to be smoothed with large HV capacitors, or a benefit in generators designed to emphasis certain non-linear phenomena e.g. displacement and radiant energy experiments. Oscilloscope waveforms of the SCR drive of a MOT, and for more details on using a MOT as a high voltage transformer see High Voltage Supply. Overall the MOT when correctly used and setup is a very robust and high power transformer, which with cooling can run at very high output powers for sustainably long time periods. Combinations of MOTs in parallel and series can generate a wide range of high current and high voltage outputs, which is the principle I have used in this high voltage plate supply.
The three MOTs are switched independently from off to specific line supply phase (live and neutral connection to the primary) by the three toggle switches T1 to T3 on the power control front panel. The MOTs themselves are physically arranged on a nylon plastic sheet so that the MOTs cores are not electrically connected. The core of a MOT usually forms one terminal of the high voltage output, the inner end of the secondary being connected directly to the core. In this way the transformers can be isolated from each other and then connected via the patch board into different combinations of single, parallel, or series connected. Configurations of the transformers using the patch board is detailed in Figures 4, and further discussed below in that section. The patch board provides both connection of the transformers together in different configurations, and also connection of the the configured transformer set to the various internal modules of the power supply as follows:
1. The OUT+ and OUT- terminals take the raw transformer output directly to the HT output board, and allow for direct drive at the output from the transformers.
2. The RCT+ and RCT- terminals connect the transformers to the HV bridge rectifier inputs, and its outputs are connected to the HT output board.
3. The DBL+ and DBL- terminals connect the transformers to the HV level shifter inputs, and its outputs are connected to the HT output board.
There are two protection fuses at the high-side and low-side of the transformer outputs and prior to connecting to any of the internal modules or HT output board. The high-side fuse is particularly good to prevent excessive current draw through the bridge rectifier and level shifter diodes, whereas the low-side fuse is particularly good to prevent spike surges from the transformers and through the diodes, when for example a vacuum tube oscillator stops oscillating at high output power, and then suddenly restarts oscillating. Both high-side and low-side fuses are necessary to protect the supply from a range of different operating fault conditions, which is very important in extreme research and prototype operating conditions. I lost one set of bridge rectifier diodes (12 x HV diodes) before I used the high and low side protection fuses. A 2A FSD AC analogue meter is connected in series with the low-side fuse, which gives an average approximation of the secondary current being drawn from the complete transformer setup. The inter-connection of the patch board, outputs, and meter is via 4mm plugs with 20kV 16AWG wire.
High Voltage Bridge Rectifier
Fig 3.1 shows the circuit diagram for the HV Bridge Rectifier, which is mounted below the patch board on the transformer module, and shown in detail in Fig 2.22. The rectifier is nominally 40kV @ 6A and is constructed from 12 x HVP2A-20 20kV 2A diodes. The diodes are mounted directly down to the nylon transformer board and again connected to the patch board and HT output board using 20kV 16AWG wire. Whilst quite well rated for the overall performance of the plate supply, semiconductor diodes are sensitive devices and easily blown short-circuit by over-current conditions, and blown open-circuit by HV spikes, transients, and non-linear power reflections from the experiments.
To protect these diodes, we use both the high-side and low-side fuses on the patch board, and also most importantly a blocking/tank capacitor at the HT output board. This capacitor significantly helps to prevent reflected transients and non-linear voltage spikes from the experiment and generator from passing back into the power supply and causing problems for the sensitive semiconductors. Typically for many experiments using a vacuum tube generator, and when a smoothed DC plate supply is not required, I use a 25kV 25nF pulse capacitor as the block capacitor at the HT output board. For DC smoothed plate supply I tend to use two 4kV 60uF capacitors in series to create a 8kV 30uF tank reservoir. A large tank like this needs very careful connection and discharging, which is one of the primary reasons a discharge unit is included in the power supply.
Overall when used with the blocking capacitor the HV bridge rectifier is robust and reliable, and can provide sustained high output power with only moderate heating of the diodes. These rectifier diodes are also in direct line of the forced cooling between the MOTs which makes for a high power high voltage rectified and smoothed DC plate supply or HV source. I have only lost one set of diodes before I had the high and low-side protection fuses installed, when operating at almost full power input and the vacuum tubes stopped oscillating during a tuning experiment. When it started oscillating again the current surge from the transformers at an almost full input power of 5kVA blew all 12 diodes short circuit. The high and low-side fuses now provide adequate protection against this fault condition, and I usually run with protection fuse ratings between 1-3A dependent on the transformer configuration, required output power, and type of generator e.g. vacuum tube, spark gap, impulse etc.
High Voltage Level Shifter (Doubler) Board
Fig 3.3 shows the circuit diagram for the HV Level Shifter or Doubler. The large microwave oven capacitor bank and 6 HV diodes that constitute the level shifter are mounted on its own nylon board, and are shown in detail in Fig 2.15. The principle of the level shifter is that in one half cycle of the secondary output the capacitor bank is charged up to the peak potential of the half-cycle e.g. 2.1kVRMS for a single transformer, and in the second half cycle a diode is used to raise (or lower dependent on the direction of the diode) the potential on the output of the capacitor bank by the maximum potential of the second half-cycle e.g. a further 2.1kVRMS for a single transformer. The overall result for a sinusoidal primary coil line supply input is an secondary output sinusoidal that is level shifted either up or down by the maximum potential of one half cycle of the waveform.
With a positive orientated diode direction this will produce a sinusoidal from 0V to 4.2kVRMS or ~6kV peak voltage when unloaded. In other words the secondary coil output waveform is level shifted either positive or negative dependent on the diode orientation, and hence why this circuit setup is properly known as a level shifter. This circuit is often referred to as a voltage doubler, but diverges slightly from a true doubler that uses multiple diodes and produces a rectified and doubled, or tripled etc. output dependent on the number of capacitor diode stages in the voltage multiplier. In this power supply I use the diode in the positive orientation to produce a positive level shifted output which can be selected using DBL+ and DBL- on the transformer and HT output boards. It is not without a sense of irony that I refer to the terminals as “DBL” or short for doubler!
The capacitor bank is an MMC type arrangement that consists of many microwave oven capacitors combined together to produce a higher capacity capacitor, and at a higher voltage. In this case I am using a bank of 3 x 1.05uF 2.1kVRMS capacitors in series to give a 0.35uF 6.3kVRMS single bank. With 11 of these banks combined in parallel the final capacitor bank is ~ 3.8uF @ 6.3kVRMS. When used in a level shifter configuration as shown in the circuit diagram this capacitor bank with 3 series input transformers can give a measured total level shifted output potential of up to 13kV @ 300mA, 15kV @ 150mA or almost 18kV peak open circuit potential. The diodes are again the same as those used in HV bridge rectifier and are arranged in 2 series banks of 3 in parallel to provide a 40kV 6A level shift diode.
High Voltage Monitor Board and Panel
Fig 3.1 shows the circuit diagram for the HV Output Monitor Board (HVOM), which is designed to safely provide a measure of the HT Output on the Power Output Monitor Front Panel, also shown in the circuit diagram. The HVOM circuit uses a HV half-wave rectifier using 2 x HVP2A-20 in series making a 40kV 2A rectifier diode. The rectified waveform is smoothed by an HV capacitor bank of 2 series and 2 parallel capacitors to form a 10nF 40kV smoothing capacitor bank. The rectifier and smoothing capacitor together turn the output waveform into a peak DC level which will be displayed on the front panel V-OUT meter as shown in Fig 2.6. The high voltage peak DC level is converted to a low current by a long series resistor chain, where each resistor is 1MΩ 2W. 20 series resistors together form the highest 20kV range and reduce the current from the rectifier to 1mA for 20kV. This dramatic reduction in current reduces the ripple on the peak DC to a very low level, and also safely converts the HT to a low current that can be passed to a meter on the front panel.
The meter on the front panel is a 1mA FSD DC analogue meter with its range updated to show kV rather than mA. So on the highest range 20kV at the HT Output Board is converted to 1mA and moves the meter needle to full-scale deflection. The 5kV and 10kV ranges are arranged by taking a tap point off of the resistor chain after 5 and 10 resistors respectively. The tap connections are arranged by a pair of HV vacuum relays which are switched by the 24V low voltage rotary position switch on the front panel. Although rated to only 3kV 10A each in this setup the relays can withstand much higher potential difference across their contacts as the current in the series resistor chain reduces the discharge current to a very low level, and hence breakdown across the contacts is suppressed.
In this way the HVOM can safely and effectively measure peak voltages up to 20kV DC in 3 ranges, 5kV, 10kV, 20kV which can be selected and displayed at the front-panel, without any HV present at the front panel controls. For additional protection from an unknown fault condition the rotary selection switch and knob on the front panel is entirely of plastic case and shaft design. It is worth noting that both the 5kV and 10kV range require one of the vacuum relays to be energised, and hence a 24V supply must be present for these two ranges. In the event of a power outage to the unit both relays will be off, and the meter will fall-back to the 20kV range by default. This must be considered carefully when using large tank capacitors which are highly charged by the supply, and they are being monitored on the 5kV or 10kV range, and then a sudden fault condition where to remove the line supply input, the meter would fallback to the 20kV range appearing to show considerably less voltage on the HV capacitor bank.
In the design of a high power, high voltage power supply it is important at the early design phase to allow for unknown and unusual fault conditions and how to protect both the operator and components from exposure to unsafe conditions. High voltage has an uncanny knack of finding the most surprising discharge and breakdown channels, and hence distance between high voltage components, breakdown resistance of insulators, and mounting materials must all be carefully considered and arranged. In this power supply all the HV components are mounted on nylon boards and supports fully isolating them from the varnished wooden casing, and from other metal and conductive brackets, mounts, and modules used in the supply construction. HV is passed around the supply on the inside using 20kV silicone coated 16AWG multi-stranded hookup wire, and the layout of the modules are so arranged to minimise the wiring length between HV modules and the HT Output board.
The inputs to the HVOM are further protected by two 1A line fuses on the low and high-side inputs. These are arranged to prevent fault conditions from destroying the rectifier, capacitors, and other monitor components in the event of an unusual fault condition in the HVOM board or monitor panel components. This was added to the design after the early prototype was being run in 10kV maximum power output test, and with a lower rated smoothing capacitor, which failed short-circuit and pulled an enormous discharge current through the rectifiers, super-heating them to a point where they exploded sending Bakelite shrapnel all around the supply enclosure and into the lab, and physically puncturing two of the level shifter microwave oven capacitors in close vicinity!
The smoothing capacitors where subsequently uprated, and fuses added to prevent reoccurrence of this kind of fault. It should however be noted that if one or both of these input fuses blow then the V-OUT monitor meter will read 0V even when there may be high tension present on the HT Output Board. It should also be obvious to the reader why careful and safe testing using the remote control is a necessity when first commissioning, and whenever operating this king of of high tension supply.
High Voltage Discharge Board
Fig 3.3 shows the circuit diagram for the HV Discharge Unit, and its implementation and construction are shown in detail in Fig 2.19. The discharge unit performs a simple and yet critical safety task, which is to discharge any high voltage that is present at the HT Output panel when the transformers are turned off. This high voltage may arise from the experiment and generator or from tank/blocking capacitors attached to the output. In a research and development environment it is usual to adapt the apparatus, experiment, and method may times during operation, and this requires being able to safely work on the equipment between operation and after fault conditions, issues, or unexpected events. This requires rapid access to a safely discharged experiment system which obviously includes the power supply. The discharge unit is an effective and reliable method to discharge very large energy stored on high capacity components in the circuit.
An example of this is as follows. The plate supply was used with the Tesla coil unit featured in the Wheelwork of Nature series, which includes a vacuum tube generator based on a single GU5B class-C Armstrong oscillator. One of the variations of this experiment used an 8kV 30uF tank capacitor at the output of the HT Output board. During extreme band-edge tuning the vacuum tube stopped oscillating, and would not restart during the experiment. With the line supply turned off at the plate supply, this left the tank capacitors charged to over 6.5kV! A 30uF tank capacitor charged to 6.5kV is storing in the region of 635 Joules of energy, which at that high potential is massive.
Discharging a high voltage capacitor with this potential and energy stored on it safely is a serious task, and cannot for example be undertaken by the old screwdriver short across the terminals. Bleeder resistors mounted permanently across the capacitor terminals are of course a necessity with a HV capacitor bank, but this takes a very long time to discharge this level of stored energy. This much potential and energy is instantly lethal under any condition, and the operator does not want to be anywhere close to the experiment or power supply whilst in this charged state. This is where the HV Discharge Board is of invaluable assistance, and when operated using the remote control, a safe and quick method to discharge this high stored energy without damaging any of the components, the HV capacitors, or the operator!
The HV Discharge board is based very simply on a high power resistor chain, in this case 5 series connected 4.7kΩ 100W 2.5kV wire-wound power resistors combine to give a 23.5kΩ 500W 12.5kV power resistor. This power resistor is capable of safely discharging output potentials up to the loaded condition of 15kV @ 150mA, from 3 series connected transformers combined with the level shifter. Although the power resistor chain is nominally rated to 12.5kV the restriction of current and short discharge time constant means that 15kV is rapidly reduced below 12.5kV without adverse effects on the discharge module. In daily use the supply very rarely operates at this 15kV level and usually only with spark gaps or thyratron generators, the normal routine being from 4-10kV for most of my vacuum tube generators. The construction of the unit is compact with the HV relays closest to the HT Output board and with the power resistors also closely connected on the lower level. Overall the unit is positioned and connected very close to the source of HT to be discharged.
The power resistor chain is isolated from the output circuit using 4 series high voltage vacuum relays, 2 on the high-side and 2 on the low-side. The combined nominal isolation from 4 x 3kV 10A relays is 12kV @ 10A. These relays also operate safely at 15kV and particularly because of the current restriction due to the resistor chain. Once again in mostly normal operating from 4-10kV the entire HV Discharge Unit is operating comfortably within its maximum nominal ratings. The unit is switched both from the front panel and from the remote control and takes only seconds to discharge the example given above of a 30uF capacitor charged to 6.5kV. The 500W load consumes the 635 Joules of energy in about 3 seconds with barely detectable heating of the resistors. I usually then leave the discharge unit on whilst I am attending to the power supply or experiment before turning off before next operation. The on condition of the discharge unit is indicated by a bright red LED on the front panel to warn against transformer operation with the discharge unit turned on.
High Tension Output Panel
Fig 2.12 shows the HT Output panel in detail. The HT+ and HT- are each connected rails which form the final high voltage or high tension outputs. The various internal modules, OUT, RCT, and DBL can be connected to the output rails using HV jumpers. The left over terminals on each rail is then very convenient for the connection of the experiment, HV capacitors, measurement probes etc. The CAP1 and CAP2 connections are provided to conveniently connect series chains of HV capacitors providing safe and intermediate connection points in the chain. The output panel also has 4mm socket and heavy-duty terminal for the transformer earth to allow experiments to be referenced directly to the floated or connected transformer earth. This panel is the only one made in nylon to prevent any leakage or discharge between module terminals and outputs when used up to the maximum 18kV open circuit condition from 3 series transformers connected to the level shifter.
Figures 4 below show the example transformer connection diagrams to setup the supply into different configurations. I have selected a range of the most useful parallel and series setups, and which also configures the supply over its full range of voltage, current, and power output. The high-resolution versions can be viewed by clicking on the following links Fig 4.1, and Fig 4.2.
In the final few sections of this post we look in more detail to the internal configuration of the plate supply using the transformer patch board, and the HT output rear panel. Any configuration of this supply must consider the requirements of the generator and experiment in terms of the required maximum voltage, current , and total power both real and reactive that will be drawn from the supply under different operating conditions e.g. varying tuning, matching, and output loading. With this established then the most simple, reliable, and optimal supply configuration can be arranged by setting up correctly the internal jumpers of the supply in order to meet the output requirements.
For example in the case of the GU-5B Armstrong oscillator coil unit used in the Wheelwork of Nature series, the nominal maximum plate potential is ~5kV. The CW power rated output when suitably cooled and driven around 1-5Mc for this tube is ~2.5kW, so at 5kV and 2.5kW of power the anode current could reach as high 0.5A. Considering current surges during extreme tuning experiments the anode current could reach considerably higher levels for very short time periods. The grid bias to keep the GU-5B oscillating under these conditions will need to be in the order of ~ 100mA – 500mA and can be adjusted using the grid bias rheostat for optimum drive matching to the experiment. Taking all this into consideration 2 series transformers will reliably supply ~ 4.2kV @ 0.8A, and up to 6kV open circuit, and 2 parallel transformers combined with the level shifter would provide ~ 4.5kV @ 0.6A, and again up to 6kV open circuit. For simplicity here I would use the 2 series transformers which also gives a better current rating, and less dissipated power with fewer HV components (less to go wrong) in the overall setup.
Now empirically the GU-5B can withstand substantially higher plate voltages when the generator is driven in low duty cycle pulsed mode, or using a staccato controller in the vacuum tube cathode connection. The advantage of this extreme operating condition is that the considerably increased anode potential will also considerably increase the peak-to-peak oscillation across the primary coil, which in turn will considerably increase the voltage magnification along the secondary coil, ultimately leading to much longer discharge streamers from the top-end of the Tesla coil secondary.
Under these operating conditions the plate supply could be as high as 9kV, and this would be best supplied by 2 series transformers with the level shifter which can supply up to 9.5kV @ 0.3A. In this extreme operating condition care needs to be taken not to allow the GU-5B to stop oscillating at full input voltage and power from the plate supply, as the tube anode would then be exposed to an open circuit voltage of almost 12kV which is too high for the GU-5B under any circumstances and could easily lead to anode-grid breakdown and destruction of the vacuum tube. Extreme operating conditions such as this have to be handled extremely carefully and with experience, but are discussed here to illustrate the setup of the plate supply necessary to operate in this region.
The other important consideration for the generator and experiment is the voltage envelope or driving waveform that is provided by the plate supply. For example the characteristics of a Tesla coil can vary enormously when the generator is driven by a sinusoidal, pulsed, chopped, or rectified and smoothed high voltage waveform. A setup consideration for the plate supply is whether to drive directly with the raw transformer output, use a rectified output with or without a tank/blocking/smoothing capacitor, or an output that is a continuous sinusoidal or chopped by an SCR. My own preference for these selections are as follows, but do very much depend on the type of generator being driven e.g. spark gap or vacuum tube, and the type of Tesla coil and phenomena that the experiment is working with e.g. Tesla’s Radiant Energy and Matter, Transference of Electric Power – Part 1, Single Wire Currents etc.
1. For experiments and generators in CW mode e.g. The Wheelwork of Nature – Fractal “Fern” Discharges, and High-Efficiency Transference of Electric Power, I use the bridge rectifier module with a tank/blocking capacitor as this allows for maximum power output efficiency from utilising both half-cycles of the transformer output, and also creates a positive forward pressure or positive voltage envelope. With a large tank/smoothing capacitor this makes for a very steady DC level anode supply which will result in high currents and hence strong, hot discharge phenomena, from powerful oscillations in the primary circuit. The blocking capacitor protects the semiconductor rectifiers from spikes and reflected power surges and transients.
2. For experiments and generators using spark gaps, or vacuum tubes in pulsed mode using a staccato interrupter, or other triggered grid devices e.g. Transference of Electric Power – Part 2, I prefer to use the raw output of the transformers in either parallel or series connection. The burst nature of the output especially with the SCR power control leads to enhancement of the non-linear and impulse like phenomena, and the setup of the pulsed triggering and staccato phasing is easier when matched to a positive or negative half-cycle envelope. This configuration is very robust for extreme operating, tuning and matching, as only the transformers are exposed to the raw output. MOTs are extremely robust provided they are not allowed to excessively overheat or are exposed to excessive series connected voltages.
3. For experiments requiring very high potentials such as Thyratron pulse generators, tank capacitor charging for impulse discharge experiments e.g. Displacement of Electric Power, I use the 2 series or 3 series configuration with the level shifter. These are specialised configurations which generate very high potentials at considerable output power, and requires considerable care and experience to operate safely. I will be covering specialised Thyratron generator usage and experiments using this plate supply in subsequent posts, but is noted here for completeness of the overall operating range and characteristics of this supply.
Parallel Transformer Setup
Fig 4.1 shows the circuit diagram configurations for the transformer patch board for parallel connection of the HV transformers. All the parallel arrangements rely on the core of the transformers being connected to Trafo earth (TRAFO_E). This connects all of the bottom ends of the transformer secondaries, the cores, together. The top-end of the secondaries are also connected together via jumpers that connect to the common positive output rail. From the common positive output rail, which includes the high-end protection fuse, a jumper connects to the raw output OUT+, the HV bridge rectifier RCT+, or the level shifter DBL+. The low-side of the connected secondaries are first connected via the low-end protection fuse through the secondary AC analogue meter and then to the negative output terminal fpr the selected output OUT-, RCT-, or DBL-.
In parallel modes it is normal to connect Trafo earth to the line supply earth via the jumper on the line supply power input panel on the rear of the plate supply. This effectively grounds the cores of the transformers to line earth and would be considered the safest configuration for running the HV supply at high output powers. However, if the generator or experiment creates considerable non-linear transients or impulses these can be passed back through to the transformers, even with a large blocking capacitor, and via the core connected secondary through to the line supply earth, and hence interfere or disturb the normal operation of other unprotected electrical equipment and instruments connected to the line earth. In this case it is sometimes necessary to remove the jumper between the Trafo earth and the line supply earth, isolating the transformers from the line supply earth.
Series Transformer Setup
Fig 4.2 shows the circuit diagram configurations for the transformer patch board for series connection of the HV transformers. In the series configurations the transformers rely on the fact that the cores are floating due to physical mounting on a nylon board. So the top-end of the T1 secondary will connect to the bottom-end of the T2 secondary or the core, with the top-end of the T2 secondary forming the high-side positive output, and the core of T1 forming the low-side output. In this 2 series transformer configuration the core of T1 can safely be connected to Trafo earth and hence the line supply earth with same considerations as in the previous section. It is important to not that the core or low-side of T2 is NOT connected to the Trafo earth, as it is in series with T1 and hence the T2 core ONLY connects to the high-side output of T1. Connection to the positive output rail, high-side and low-side protection fuses, and the output modules OUT, RCT, and DBL are the same as for the parallel connections in the previous section.
The case of 3 series transformers is a special one and needs more careful consideration. When the core of a MOT is connected to line earth as it would be in its normal primary use in a microwave oven, the potential difference between the core and the primary is only the line supply voltage, and the potential difference between the core and the secondary high-end is the maximum rated output of the transformer which is normally ~ 2.1-2.3kVRMS. The normal construction of a traditional MOT makes sure that both the primary and secondary coils are adequately insulated from the core and any magnetic shunts, according to their specific purpose, which is usually accomplished with resin impregnated and sealed cardboard or a form of thin plastic insulation kept in place again with resin.
In the case of unearthed cores in series arrangements the cores are now biased to potentials well above the primary line supply, and in this case we rely on the insulation of the primary and secondary coils from the core. In a 2 series transformer arrangement at maximum output the core of T1 is at line supply earth or for example 0V which creates no problem for the T1 primary coil, and the T2 core is at ~ 2.1kV which also does not present a problem for most good condition MOTs. The high-end of the T2 secondary is then at ~ 4.2kV, the differential across T2 again only being ~ 2.1kV. In this way the 2 series transformer arrangement can be used safely and stably without breakdown between the core and the secondary, or the core and the primary. Open circuit without a load the core of T1 will be at ~ 3kV and the output at almost 6kV which is also ok for this arrangement, as MOT design covers the open circuit fault condition in a microwave oven.
This would not be the case for a 3 series transformer arrangement where T3 was simply added on top of the 2 series setup. Now the core of T3 would sit at ~ 4.2kV and the high-end of T3 at ~6.3kV, and this is under maximum output and full load. Open circuit the core of T3 would sit at ~ 6kV and the output at almost 9kV. The 4.2 – 6kV potential of the T3 core is too much potential difference between the primary coil and the core. The windings of the primary coil in T3 are still at the line supply voltage level, and most MOT insulation will fail when exposed to this 6kV potential difference, resulting in strong breakdown between the core and primary, and in some cases the secondary and core, and this is for a good condition transformer.
To use 3 series transformers safely and reliably the Trafo earth must be moved to the midpoint of T1 and T2 so that the T1 core and the T2 core are connected to Trafo earth and hence line supply earth. Now the phase of the line supply is adjusted for T1 and T2 to be in anti-phase to each other (via the front-panel transformer switches), and so the T1 high-end of the secondary goes negative -2.1kV and the high-end of T2 goes positive +2.1kV, the potential difference across the two transformer outputs being again ~4.2kV. Now T3 can be added on top of the T2 output in series with the T3 core sitting at 2.1kV fully loaded, and 3kV open circuit. The T3 phasing of the primary is set to the same as T2, and opposite to T1. Now 3 series transformers can produce a loaded output potential difference of 6.3kV, and ~ 9kV open circuit, without breakdown between the core and the primary coils at the line supply.
In relation to Trafo earth or line supply earth if connected, then T1 high-side is at -2.1kV or -3kV OC, T2 high-side is at +2.1kV or +3kV OC, and T3 high-side is at +4.2kV or +6kV OC. In this configuration the negative side output of the power supply is now NOT earth, which is very important when connecting the vacuum tube generator. The negative rail is now -2.1kV and hence both the generator and the experiment must use the negative rail as the bottom-end or base connection for the various units, and NOT the line supply earth. To connect the generator and experiment to line supply earth at the bottom-end would be to short the output of transformer T1, which will throw-out the MCB at sufficiently high input current.
Operation, Line Supply and Safety
Operation of a high voltage high power supply like this one should always be undertaken with great care and caution and with well defined method that is adhered to throughout its operation. Establishing a good operation procedure introduces a disciplined approach, and reduces the chances of unexpected events and mishaps arising from careless use. Remember that a high voltage supply is instantly lethal if not used correctly. What follows here are some of my own procedures when working with this type of high voltage supply:
1. Always where possible operate the supply using the remote control at a reasonable distance from the high voltage supply.
2. When approaching the high voltage supply always check the V-OUT meter is on zero, and if not use the discharge control on the remote control.
3. When setting up the power supply configuration using the transformer patch board, or adjusting any internal part of the supply, make sure that the line supply is turned-off at the primary line supply input panel at the rear, and that any capacitive elements in the system are discharged.
4. Always test a new configuration of the supply at very low input power, to check that setup has been accomplished correctly.
5. When tuning an experiment always run the high voltage power supply at low output power until the correct operating point has been found.
6. Wind up the power to an experiment slowly, restricting high power operating to short bursts until satisfied that the supply, generator, and experiment are stable and can withstand longer sustained high power operation.
7. For sustained high power operation turn on the cooling fans, and preferably close the supply top panel in order to improve the cooling efficiency. During long periods of experimentation at high-power allow the system to cool intermittently, and do not allow the transformers cores to become overheated.
8. If a protection fuse is blown, disconnect the generator and experiment and safely investigate the reason and source of the fault event.
9. Never rush to change the power supply setup, and never leave the power supply operating unattended.
10. Arrange if possible a single master emergency power-off switch which will cut all power to the supply, and if the experiment produces phenomena with strong dielectric and magnetic induction fields consider wearing appropriate protective gear.
Adhering to these kind of safety procuedures in setup and operation are critical when working in high voltage research and development. The supply presented here is robust, and with a very wide range of output performance, and when used safely and correctly with suitable generators and experiments, is capable of covering the wide range of phenomena generally accessible in the alternative electricity research field, with power levels up to 5kW.
Another module in the tube power supply series is a heavy-duty line supply filter and power factor correction unit. This module which attaches between the line supply and power input rear panel, performs two important jobs. The first is to isolate the line supply from higher frequency transient noise coming back through the experiment, generator, and plate supply, which is important if the experimental apparatus is setup in a domestic setting, or close to any other more sensitive electrical equipment such as computers and digital communications equipment etc. My research lab is arranged in a rural industrial environment that caters for a lot of welding, and other electrical disturbance processes and apparatus, and hence the short run electrical disturbance created by my Tesla coil experiments does not disturb other endeavours or power supply users.
The second job is to correct the low power factor that arises from running microwave oven transformers. The very high inductive load of a MOT, and especially multiple MOTs driven either in parallel or series configurations easily reduce the power factor to ~ 0.6 or even down as low as ~ 0.4. This is not ideal for longer term high power experiments where the input currents can become very high and the overall apparent input powers can rise as high 10kVA when using all three transformers flat-out. Power correction using parallel connected PFC capacitors is accomplished by this module and uses a range of jumper selectable capacitors to improve the power factor during longer experimental runs. Overall the actual running time of a Tesla coil in a research environment is usually limited to short run bursts, and hence the impact on the line supply in the correct industrial setting is minimal. This line supply module will be covered in detail in a subsequent post.
Overall the 5kW high voltage and plate supply presented in this post is very robust, is easily configured to a wide range of different output voltage and power levels, and is also relatively straightforward to operate with the necessary experience and know-how. This supply is intended for an electricity research and development environment using Tesla coils and associated generators, and in a non-commercial and non-industrial setting. This power supply will feature in quite a few of the experiments yet to be presented on this website, which will also show more detail as to the setup, usage, and operating characteristics of the complete tube power supply series.
The next parts in this tube power supply series will cover the individual tube board designs and configurations for parallel and push-pull tube operation, and also pulsed power using a staccato interrupter.
1. A & P Electronic Media, AMInnovations by Adrian Marsh, 2019, EMediaPress
2. Dollard, E. and Energetic Forum Members, Energetic Forum, 2008 onwards.
In research using Tesla coils it is inevitable that sooner or later a vacuum tube power supply will become a necessary and invaluable addition to the laboratory equipment. Vacuum tubes when correctly setup and operated are a robust and high power solution to driving Tesla coils from very low frequencies, and to well into the HF frequency band. Most of my experiments are conducted in the 160m amateur band with a centre frequency around 2Mc, and with tuning that can go down as low as 500kc, and up to almost 4Mc. A vacuum tube generator that can be flexibly configured to drive different configurations and types of tubes to power levels over 1kW, and even up to as high as 5kW, opens the door to many fascinating and unusual electrical phenomena, that can be observed and measured using Tesla coils driven at higher powers and higher frequencies. This post is the first in a sequence to look at my own tube power supply, designed specifically with rapid prototyping and Tesla coil research in mind, and is the product of using vacuum tubes of various different types and configurations in my research over the years.
Note: A high voltage supply is capable of delivering voltages and currents, even at lower powers, that are instantly lethal, and that any design and operation of a high voltage unit should be undertaken with great care by a trained and experienced individual. I have so far presented on my website a basic and yet configurable Vacuum Tube Generator based around dual 811A’s, and which has been used in a range of already reported experiments including, Transference of Electric Power, Single Wire Currents, and Tesla’s radiant energy and matter. In this post I start looking at a much more comprehensive tube power supply that I use on a daily basis with a range of different tube boards. I will be looking at the design, construction and operation of the heater, grid & screen supply (TPS-HGS), including a video overview and simple experimental demonstration of its basic operation. More detailed and sophisticated operation will be covered in subsequent experimental posts as I publish them.
Before launching into the details of this supply, I will first give an overview of my complete tube power supply system, and its major components:
1. The heater, grid & screen supply is covered in this post, and provides the filament heater supply to the installed tube board with variable control up to a maximum 12.6V @ 25A, a finely controllable grid bias supply with wide operating characteristics between ±750V DC @ 200mA, or a finely controllable screen or auxiliary bias supply up to 1500V DC @200mA.
2. A high power 5kW plate supply using three 1.8kVA industrial microwave oven transformers, that can be configured in a variety of parallel and series arrangements to provide plate supplies including 2kV @ 2.3A, 4kV @ 0.8A, and up to 6kV @ 0.8A. A high voltage 40kV 6A bridge rectifier is incorporated into the design, along with a 12kV rapid discharge unit for safely discharging tank capacitors in the driven circuit. Also internally installed is a 4uF 6kV level shifter to increase the output up to 12kV @ 300mA and 15kV @ 150mA, which is suitable to drive medium power thyratron tubes, such as the 5C22 for pulse and impulse discharge experiments, as well as displacement of electric power experiments. I will be covering the design, construction and operation of this supply in a subsequent post.
3. A dual 833C RF Power Triode tube board with graphite plates and with continuous axial cooling, driven at 4kV plate supply and with a total usable output power of ~ 3.0kW @ 2Mc, and the heater drive is 10V @ 20A AC for both tubes. The graphite plates of the C variant of the 833 tube improve significantly the top-end performance of this tube board by reducing plate to grid flash-over under high-power or poorly matched output conditions. Suitable for displacement and transference of electric power experiments, Tesla’s radiant energy and matter experiments, and including plasma, induction generator, and discharge phenomena.
4. A quad 811A RF Power Triode tube board with continuous axial cooling, driven at 1.2kV plate supply and with a useable output power of ~ 1kW @ 2Mc, and the heater drive is 6.3V @ 16A for all four tubes. This is a very versatile and flexible day-to-day workhorse with lower plate supply requirements, and facilitates a wide range of Tesla experiments as already demonstrated on my website using power up to 1kW.
5. A dual 4-400A RF Power Tetrode tube board with continuous axial cooling, and which is particularly good for high-fidelity musical Tesla coils, and linear amplifier type experiments where modulation and signal purity combined with good output power are required.
6. A dual 810C Power Triode tube board with graphite plates and continuous axial cooling, and which is particularly good for driving lower frequency Tesla coils in the hundreds of kilocycle frequency range, and with good power modulation and signal linearity.
The design, construction and operation of these tube boards will be covered in more detail in subsequent posts, and also operation of the complete tube power supply system as part of experiments yet to be presented on the website. So let us now get on with the tube power supply – heater, grid & screen unit with a video overview of its design, construction, and operation, and including driving a basic experiment using a single cylindrical Tesla coil with a single wire load. The video also demonstrates the use of both the dual 833C and quad 811A tube boards, here used as tuned plate class-C Armstrong oscillators, deriving linear feedback directly from the secondary coil oscillation, and primary circuit tuned to drive the cylindrical Tesla coil at the upper and lower parallel resonant frequencies.
The principle of operation for the heater supply unit is as follows. This supply provides a high current low voltage output to drive the filaments in the tube board when connected in series or parallel arrangements. The internal resistance of the vacuum tube filaments determine the supply requirements without any additional regulation at the supply end. To this effect a 12V 300VA transformer can be adjusted using a variac to correctly bias the requirements of the tube board both in voltage and current. The power rating of the transformer was selected to adequately cover the various tube boards being used, and is capable of a maximum of 12.6V @ 25A. Open circuit the supply provides 15.9V which reduces with increased load, and to the correct filament voltage and current when adjusted by the variac.
A soft-start switch is incorporated to switch a resistive load 50Ω 50W into the primary circuit of the transformer, which reduces the potential across the primary, and hence reduces the secondary output. When vacuum tubes are cold the filament resistance is generally much lower than when in normal operation, and the initial in-rush of current when power is first applied to the filament circuit can easily exceed the maximum safe ratings, which can lead to significantly reduced filament lifetime and premature failure of the filaments in one or more tubes. The ac voltage and current supplied by the heater supply is monitored using an analogue true rms circuit through a DC 1mA ammeter, and a digital 50A AC ammeter based on the potential difference across a 75mΩ series resistance in the output circuit.
The digital ammeter is most effective for setting accurate bias current prior to RF circuit operation. The outputs of the heater circuit are arranged flexibly on the back panel to allow rapid and configurable connection to the tube boards, and including the ability to float the filament supply above the line supply earth. Disconnecting the heater supply from the line earth allows the vacuum tube to be cathode switched, modulated, or “pulsed”, and for the tube board to be referenced to a different “ground” e.g. a dedicated RF ground, or plate supply with high voltage biased negative, (useful for extreme high-voltage thyratron supplies).
The principle of operation for the grid/screen bias unit is as follows. This supply provides a stable unregulated output bias based on the voltage accumulated in a tank circuit, and which can be finely controlled by a high power potential divider to the output. A high-voltage transformer with dual secondary coils rated at 500V each with a total power output of 250VA is adjusted using a variac on its primary circuit. This gives a variable output voltage of ±500VRMS @ 200mA when negative reference is at the centre tap, or 1000VRMS @ 200mA when negative reference is at the bottom-end of the lower secondary coil. The output of the high-voltage transformer is bridge rectified and then accumulated on a tank capacitor circuit consisting of 4 x 560µF 450V capacitors in series. Bleed resistors and a high-power parallel load resistance are provided for rapid discharge of the tank when switched off. The tank is intended to provide a stable DC supply with very low output ripple up to 200mA for grid and screen bias purposes.
To facilitate very fine adjustments in grid bias, which is often very necessary to establish the best operating point for a tube amplifier or oscillator, the output of the tank circuit is fed through a 150W 10kΩ rheostat, which provides continuous linear adjustment of the output across the entire range of the tank voltage. This allows for initial setup of the tube board prior to application of the plate supply, and then variable bias tuning during operation of the experiment. As the bias output is unregulated changes to the experimental conditions will effect required changes to the grid bias and this can be safely and readily applied through the grid bias rheostat. The final result is a very flexible supply that can accommodate a wide range of different tubes and operating conditions. Rapid adjustment of these parameters in a research and development context greatly reduces experimental setup and adjustment time, and facilitates easy tuning to find the most optimum point of operation.
The rheostat fine control is fed from the tank capacitors via two changeover high-voltage relays that switch the output between the upper and lower secondary coils, or across both coils. This allows the output range to be more precisely and safely controlled by selecting just a negative output range, a positive output range, or the entire tank range. This has benefit for example when biasing a tube board in grounded cathode for linear amplifier application. Here the grid bias for a class C linear amplifier is usually in the negative range, so to minimise power dissipation in the grid adjust rheostat, and to ensure that the bias cannot drift into positive voltage with higher risk of tube damage, the output relays are configured to connect only the negative section of the tank circuit across the grid adjust rheostat and hence to the output.
Measurement of DC tank voltage, and output bias voltage is accomplished by a switched series resistance which scales the current into an ammeter up to 1mA. For greater accuracy and scale size the analogue meter is switched either to measure a negative bias potential, or a positive bias potential, by switched reversal of the measurement current through the meter. This series resistance method gives a very good dynamic range of measurement with ranges between 20V DC FSD, and 2kV DC FSD. The process of operating the grid/screen supply requires that the tank voltage first be set to a value higher than is required for the output bias, and then the output bias set through the fine control of the grid adjust.
The switching between these measurements is quickly and easily facilitated by the rotary controls on the front panel of the instrument. The rotary switches are plastic spindle types, which also provide excellent isolation during operation from internal high voltage. It should also be noted that the switched series resistance also has part of that resistance chain in the negative terminal of the output e.g. R14, R15, R16, R17. These resistors prevent current surges between the various output circuits during switching of the measurement ranges, and also inadvertent changes to the setting of the tank output relays when the tank circuit is not discharged. This is particularly important at high tank voltages where switching could otherwise result in large surge currents and destruction of the relays, and other switching components. I discovered this one during inital supply tests, and needed to change both relays and a rotary switch that had burned and fused contacts from a surge at maximum tank setting of 1500V!
Measurement of DC output current is by digital ammeter with 200mA FS. The digital meter is a 200mV FS DC meter which has a 1Ω 2W shunt resistor at the output of the grid adjust rheostat. As for the heater supply, the digital readout facilitates accurate bias adjustment and setup prior to operation of the tube board at RF frequencies. Overall the two supply units are simple in design and construction, and compact and cost effective in materials and components, but lead to a very wide range of operating characteristics, which can be quickly and easily adjusted by a skilled operator during the experimental process.
Figures 2 below show the complete unit with both the dual 833C and quad 811A tube boards installed. The pictures illustrate the compact yet powerful design, and particularly the space saving footprint on the bench. When combined with the 5kW high-power plate supply, the two together form a very versatile and robust tube power supply suitable for a very wide range of Tesla and high-voltage research experiments including, displacement and transference of electric power, Telluric transmission of power, radiant energy and matter, modulated and high-fidelity waveforms, and plasma and discharge phenomena. The same plate supply combined with a specialised 5C22 thyratron board and pulse trigger unit is well suited to displacement of electric power, pulse, impulse, and unidirectional discharge phenomena.
Figures 3 below show the internal layout and construction of the complete heater, grid & screen tube supply. The entire unit is housed in an oil varnished plywood housing, with consideration for cooling, correct line earthing of the appropriate components, internal safety of the high-voltage components and regions, and the external safety of the operator with the various controls when adjusted during the experimental process. As discussed on the video, the choice of a wooden enclosure faciltates easy fabrication and construction, with reasonable thermal properties when fan-cooled, and reasonable external isolation from high-voltage components and regions.
The wooden enclosure does not facilitate grounding and earth connection of certain components, which requires more considered wiring and interconnection of line earth around the internal layout. The wooden enclosure provides no EMI protection either externally to other objects in the facinity, or internally from electric and dielectric fields of induction around the experiment. In a research and development environment in an industrial and isolated setting this is considered acceptable given the often short operation time periods, and minimum interference to surrounding infrastructure.
It can be seen from figures 3 that the overall layout and construction is relatively straightforward. Care with proper positioning and wiring of the high voltage components is very important, particularly in spacing of contacts, the wire type used to connect the high voltage components, and isolation from the user controls on the front-panel. Otherwise a flexible design is possible from a simple circuit, is easy to diagnose and fix if and when a problem occurs, and facilitates a very wide range of experimental conditions that can be adapted, adjusted, and tuned quickly in a research and development prototype setting.
The next parts in this tube power supply series will cover the plate supply, and the individual tube board designs and circuit configurations.
1. A & P Electronic Media, AMInnovations by Adrian Marsh, 2019, EMediaPress
2. Dollard, E. and Energetic Forum Members, Energetic Forum, 2008 onwards.
In this post the cylindrical coil transmission gain S21 is explored using the DG8SAQ vector network analyser. The small signal ac input impedance Z11 has been explored and presented extensively for both flat and cylindrical Tesla coils, and the transmission gain study in this experimental post continues the small signal analysis of this type of Tesla coil. The S21 characteristics show that the Tesla coil has its lowest insertion loss at the fundamental series resonant frequency, and its highest loss at a parallel mode. The series resonant mode remains relatively stable with changing primary tuning characteristics such as number of turns, and variations in the primary tuning capacitor. However, the parallel mode shows strong dependence on both the primary turns and primary tuning capacitor.
A Tesla coil is a passive network element in that it has no active power supply and hence no power gain, so in transmission gain measurements we would expect the maximum gain or minimum insertion loss to be 0 dB in theory. In practise of course there are always losses introduced through various aspects of the system, and the maximum gain will be less than the ideal 0 dB. It is important here to distinguish the difference between power gain and voltage magnification. A vector network analyser measures the reflected and transmitted power between its input and output ports, and hence the resulting scattering matrix reveals the characteristics of the network based on the proportion of power incident at each port. With a suitable calibration this scattering matrix can be converted to a range of different parameters including impedance, component values, as well as gain and loss. So in measuring a Tesla coil as a passive network element the transmission gain will always be less than 0 dB.
The Tesla coil through induction field coupling from turn to turn introduces voltage magnification and charge accumulation across the turns of its secondary coil. This is accompanied by the appropriate reduction of current in the secondary, so the overall power gain of the system remains less than 0 dB. The resulting magnification of the secondary can generate very high potentials at the top-end of the secondary coil, and when combined with a suitable capacitive top-load, accumulation of significant energy when pumped by successive cycles of the generator in the primary. The high tension at the top-end combined with the accumulated stored energy can lead to very significant and spectacular discharges, which in themselves often reflect core qualities of the Tesla coil type and geometry, as well as the type of power supply and operating characteristics (frequency, modulation etc.). A great deal of research and investigation into the underlying nature of electricity is possible by working directly with a Tesla coil that has sufficient magnification to produce a discharge at its top-end, or pump significant power into a single wire transmission medium at its bottom-end.
At first order the transmission gain characteristics of a Tesla coil present as a high-Q bandpass filter typical for a resonant circuit, and where the insertion loss for a direct connected secondary coil is in the region of 4-5 dB at the fundamental series resonant frequency. Direct connection of a secondary coil to the measurement equipment introduces loading to the coil which substantially changes the free resonant frequency of the coil, shifting it downwards by up to ~ 1Mc. In order to measure the free resonant characteristics of the Tesla coil in transmission mode it is more useful to place the output probe a small distance ~ 2-5mm from the top-end conductor, forming a capacitive pickup to the top-end output of the coil. This allows the coil to more freely resonate according to its intrinsic characteristics, but does introduce an additional insertion loss, according the capacitive connection to the probe at the frequency of operation. In this the case the capacitive probe is only 1-2pF which introduces ~ 20dB insertion loss @ 2Mc into the measured results.
At second order the transmission gain characteristics of a Tesla coil present a wealth of interesting detail and phenomena. In this post we explore the S21 characteristics of a cylindrical Tesla coil using the measurement process thus described, compare and contrast the results to the simultaneously measured Z11 input impedance characteristics, and look at the dependence of the transmission gain to different circuit elements, including primary tuning and magnetic coupling coefficient. We also look at an equivalent circuit model that yields well matched theoretical characteristics to those measured, and which assists in understanding the mechanisms contributing to the unusual and fascinating characteristics of the Tesla coil.
The video experiment demonstrates and includes aspects of the following:
1. The experimental setup using the DG8SAQ vector network analyser for transmission gain measurements S21 for a cylindrical Tesla coil.
2. The characteristics of S21 and S11 when the primary tuning capacitor is set to balance the parallel modes on the measured input impedance Z11.
3. The changing characteristics of S21 and S11 when the primary tuning capacitor is adjusted through its full range of 20pF – 1280pF.
4. The changing characteristics of S21 and S11 when the number of primary turns is varied between 1 and 4.
5. The changing characteristics of S21 and S11 when the distance between the primary and secondary coil is varied from 7cm up to 75cm.
6. The series and parallel resonant modes revealed in the transmission gain S21, and their variation dependent on the interaction between, and the electrical characteristics of, the primary and secondary coils.
Video Notes: For clear viewing and reading of the VNWA software measurements, “720p” or “1080p” video quality is recommended, and may need to be selected manually from the settings icon once playback has started.
Figures 1 below show the key measured S21 and Z11 small signal characteristics presented in the video experiment, along with a more detailed analysis and consideration as to their possible origin and effects on the overall properties of the TC. In the presented measurements S21 and Z11 can be identified as follows:
Blue – S21 magnitude in dB, scale 10dB/div, and 0dB reference level at the top of the vertical axis.
Red – S21 phase in degrees, scale 90°/div, and 0° reference at the vertical axis centre line.
Orange – Z11 (from S11) magnitude of input impedance in ohms, scale varies but default is 2500Ω/div, and 0Ω reference at the bottom of the vertical axis.
Green – Z11 (from S11) phase in degrees, scale 90°/div, and 0° reference at the vertical axis centre line.
To view the large images in a new window whilst reading the explanations click on the figure numbers below.
Fig 1.1. Here we see the basic form of the transmission gain S21 when the return probe is connected directly to the top-end final copper turn of the secondary coil. The secondary coil is of course loaded by the 50Ω input impedance of the VNWA which causes the free resonance of the secondary coil, (nominally 1.95Mc in the 160m amateur band with the bottom-end rf grounded, and ~2.25Mc with a 2m wire extension at the bottom-end), to be dramatically reduced in frequency to M2 @ 1.45Mc and with an insertion loss across the complete system of 5dB. Calibration of the VNWA confirmed that insertion loss without the TC was << 0.1 dB.
The form of the transmission gain takes on that typical for a high-Q resonant circuit where at the series fundamental resonant frequency the gain peaks with a quality factor determined by the resistive losses in the system which is dominated here by the series resistance of the secondary coil. From the input impedance characteristics Z11 @ M2 we can see the transformed down series resistance of the secondary coil in the primary RS is 25.8Ω. The phase of the transmission gain around the series resonance at M2 is also typical for the characteristics of the series resonant TC and represents the transition of the secondary from an inductive element to a capacitive element with the corresponding phase shift from +90° to -90°.
In correspondence the characteristics of Z11, here shown in the unbalanced parallel mode condition, shows the fundamental series resonant mode with minimum series resistance in the primary at M3 @ 1.47Mc with RS = 10.4Ω. The correspondence of M2 and M3 is very close here, but not exact. This results from the tuning in the primary which in this case is a very unbalanced condition for the parallel modes in the primary and secondary at M1 and M4. In this case the upper parallel mode at M4 from the primary dominates, which is more than sufficient to skew the characteristics of the secondary coil when coupled to the primary in an unbalanced fashion as demonstrated. In the transmission gain markers at M2 and M3 can be seen to be very close but not exact.
This slight mismatch of the series mode in the primary and secondary would appear to be insignificant, but does lead to interference in the cavity when trying to tune for very high efficiency of transference of electric power in a TMT cavity, and hence instability and loss of selectivity in the tuning process, making it very difficult to sustain the highest efficiency transference of power. Where possible maintaining the parallel modes in optimal balance considerably reduces this instability and facilitates tuning a TMT system to stably transfer high power in a sustained fashion with efficiencies > 99% in the close mid-field region.
Fig 1.2. Shows the effect of moving the return probe from the top-end of the secondary coil to the closely spaced plastic guard ring, shown on the video above the copper shield turn. Removing the loading from the top-end of the secondary coil allows to freely resonant according to its intrinsic properties, and reveals a most interesting second order effect in the overall properties of a TC. The transmission gain S21 now demonstrates both a transmission peak from the fundamental series resonant mode at M2, and a parallel resonant mode at M4 where the impedance of the secondary effectively becomes very high, and no power is transmitted through the coil, rather being stored in the coil instead at this frequency. This parallel mode can be identified as properly a second resonant mode of the coil based on the sharp phase change occurring at M4. In figures 3, later in this post, we look at a simple equivalent circuit model for the TC resonant circuit that demonstrates how this characteristic of a series and parallel mode may arise.
Before going there if we look in more detail at the measured S21 characteristics. Firstly for the series fundamental resonant mode at M2, the maximum transmission gain has shifted back up to that expected for the coil design in the 160m amateur band and bottom-end connected by a 2m extension wire. This series resonance occurs at 2.27Mc and is the minimum input impedance drive point transformed down into the primary RS = 3.9Ω. This input series resistance in the primary is properly the combination of the resistance presented by the primary circuit and the transformed down series resistance of the secondary coil at resonance. This impedance transformation into the primary is based on the square of the ratio of the secondary to primary coil turns, and then scaled by the magnetic coupling coefficient k. If we assume the series resistance of the primary circuit to be exceedingly low << 0.1Ω, then the measured value for RS of 3.9Ω is entirely from the transformed down secondary coil:
Impedance transformation of the TC based on the turns ratio = (NS/NP)2 = (24/3)2 = 64
Series resistance of the secondary at M2 the fundamental series resonant frequency of 2.27Mc = 3.9Ω x 64 x k ~ 67Ω, (where the magnetic coupling coefficient k was determined empirically to be ~ 0.27).
The insertion loss of this series mode has now increased significantly from 5dB to 23.9dB based on moving the return probe from direct circuit connection to capacitive connection at the top-end of the coil. This capacitive connection of 1-2pF introduces an ~ 20dB loss in the transmitted signal that remains constant throughout the rest of the measurements. Empirically we find that the overall insertion loss of the TC, factoring in the loss from the probe proximity, to not have changed significantly and is of the order of 4-5dB. The capacitive probe coupling is an order of magnitude less than the self-capacitance of the TC system, and hence is not expected to substantially influence the measured form of the S21 and Z11 characteristics over the measured band.
By unloading the top-end of the coil and allowing the secondary to freely resonate we have revealed a most important second order effect that relates to a parallel vibration mode in the secondary coil, conjectured to arise from the distributed inter-turn capacitance from the geometry of the coil, and conjectured to instigate the formation of a longitudinal magneto-dielectric transmission mode (LMD), in the electrical cavity of the secondary coil and its extension. In this case the parallel mode at M4 is at a frequency of 3.33Mc and is a real resistance maximum where energy is not transmitted through the TC, but can be stored or accumulated in the coil, and particularly in a top-load if one where connected at the top-end of the secondary coil.
It should also be noted that the parallel modes measured in the input impedance characteristics Z11 have been balanced by adjustment of the parallel tuning capacitor in the primary CP = 552.3pF. The lower parallel mode is from the secondary at M1 = 1.95Mc, and the upper parallel mode is from the primary M3 = 2.71Mc. It remains to be determined if and how the lower and upper parallel modes measured in the input impedance correlate with the parallel resonance mode in the transmission gain secondary. Some consideration of this will be made in the following measurements looking at the dependence, of the both the series and parallel resonant points in the transmission gain, on configurable parameters of the TC system such as the number of primary turns, and the coupling with distance of the primary and secondary coils.
Fig 1.3. Here the primary tuning capacitor has been adjusted to be fully open at its minimum value of CP = 18.6pF. This significantly unbalances the parallel modes in the input impedance Z11 so that the parallel mode of the secondary is now at M1, and the mode from the primary has now shifted off the top of the measured band >> 5Mc. The series mode in the transmission gain, both in frequency and insertion loss, is only very slightly effected to 2.25Mc and 22.8dB. It should be noted that the large imbalance on the input parallel modes introduces a slight misalignment of the series mode in the input and the series mode in the transmission gain, which can be seen in the difference between markers M2 and M3, a difference of ~ 20kc. Interestingly the parallel mode in the transmission gain also remains reasonably constant at M4 3.37Mc, up from 3.3Mc in the balanced condition, a difference of 40kc.
A linear amplifier oscillator would be best tuned to the series mode at M3 for maximum transference of electric power through the TC or TMT system. Although drive point at M2 has a very slightly lower insertion loss in the transmission gain, the input impedance at this point is significantly more than for M3. At M3 the input impedance is purely resistive and represents the best match to the generator in transferring power from the generator to the primary circuit, whereas the impedance at M2 is higher and has an associated reactance, so not a pure resistive impedance at resonance.
Fig 1.4. Here the primary tuning capacitor has been adjusted to be fully closed at its maximum value of CP = 1280.2pF. This again significantly unbalances the parallel modes in the input impedance Z11 so that the parallel mode from the primary now dominates at M1 1.48Mc, and the parallel mode from the secondary is now pushed to the upper mode and heavily suppressed at M4 2.37Mc. Once again the series and parallel mode transmission gain characteristics are only very slightly affected moving no more than 20kc from the balanced condition. It should be noted that the optimal series primary mode drive point has now shifted down to M2, and away from M3 as per the previous minimum CP tuning in Fig. 1.3. The stable drive point for a series feedback oscillator would now be at M1 1.48Mc.
Overall the last two figures have looked at the impact on the transmission gain of the TC by tuning the primary tuning capacitor through it maximum range. It can be seen from the measurements that whilst this has significant import on the input impedance of the TC system, and hence the optimum drive points for different types of generators, it makes only the smallest difference to the series and parallel resonant modes in the secondary coil. This relative independence between the matching and tuning of the primary and secondary modes of the TC, has been well utilised in the Transference of Electric Power experiments, in order to tune the TEM mode for maximum power transfer from the generator to a TMT cavity, and then for LMD mode tuning in the cavity of the TMT between the two TC endpoints. The overall result when both the TEM and LMD modes are tuned optimally in the complete TMT system, is high-efficiency transference of electric power down a single wire transmission medium in the mid-field region, explored and reported so far in High-Efficiency Transference of Electric Power parts 1 and 2.
Fig 1.5. In the next two figures we look at the changes in the transmission gain characteristics with changing number of turns in the primary. Here the primary windings have been increased from 3 to 4 turns, and the TC has been tuned using the primary tuning capacitor to balance the parallel modes in the input impedance Z11. The effect on the series resonant mode in transmission gain S21 is only slight, with the frequency remaining almost completely constant at M2, 2.27Mc. The increased magnetic coupling from an extra turn has reduced the insertion loss from 23.9dB to 21.9dB at M2. The increased magnetic induction field coupling has also intensified the lower and upper parallel modes in the input impedance shifting the peaks to higher impedance, and hence a vertical axis scale shift from 2500Ω/div to 3500Ω/div. However the most remarkable change is in the parallel resonant mode in S21 which has shifted dramatically down in frequency from 3.33Mc in Fig. 1.2 to 2.99Mc, a shift of 340kc.
From our previous discussion we have so far considered the possibility that this parallel resonant mode in S21, that may originate from the distributed inter-turn capacitance of the secondary, is also strongly affected by the distributed capacitance in the primary as well. This leads me to conjecture that the parallel resonant mode in the transmission gain is influenced by the extension of the dielectric induction field from the primary to the secondary, or a capacitive coupling across the turns of the primary and the secondary coil together. If this were the case it would give a more complete view to the transference of electric power across an entire TMT system, and thus far explored in the research currently presented on my website.
For power to be coupled from the generator and through a TMT system via a single wire or Telluric transmission medium to a distant load, it is necessary for the dielectric and magnetic fields of induction to be transferred from source to load, or to extend, albeit in this case incoherently, across the complete system. Power transfer in this regime through induction in a TC requires both the dielectric field extending across the inter-turn distributed capacitance of the primary and the secondary, whilst the magnetic field is coupled between the primary and the secondary coils. Together both induction fields lead to a balanced and equilibrium circuit condition that requires both the TEM arrangement in the primary of the transmitter and receiver TCs, and the LMD mode in the single wire medium of the cavity between the secondary end-points.
Whilst this is purely a conjecture at this time, and relies both on the LMD transmission mode model, and induction field mechanics in the TC transformers, it does appear to me as an interesting and consistent expression of the balance and cooperation required within the inter-dependent relationship formed between the differentiated induction fields at the level of transference. We will see further in figures 3 how the parallel resonant mode in S21 varies strongly according to the distance between the primary and secondary coils, additionally suggesting dielectric induction field continuity between the two coils in the TC system.
Fig 1.6. Here the number of primary windings have been reduced from 3 to 2 and the input impedance rebalanced. The reduction in the magnetic field induction is clear to see in the transmission gain S21. At the series resonant point at M2 2.24Mc, the insertion loss has now increased from 23.9dB to 25.3dB, and the parallel modes in the input impedance characteristics have been reduced as there is reduced interaction between the parallel mode from the secondary and the parallel mode in the primary, (the vertical scale back to 2500Ω/div, and a reduction in parallel mode peak height from Fig. 1.2). The parallel resonant mode of the secondary has remained relatively constant with Fig. 1.2 only having reduced slightly from 3.33Mc to 3.29Mc.
Figures 2 below build upon what has been explored so far, and looks at the transmission gain S21, and the input impedance Z11, as a function of the distance between the primary and secondary coils, and hence on the dielectric and magnetic induction field coupling and continuity between the two coils.
Figs 2.1-2.5. The progression of the coil characteristics over the first 5 figures spans a primary to secondary coil distance from 7cm up to 40cm. The two coils that constitute the TC are moved progressively out of proximity with each other reducing the magnetic and dielectric induction field coupling between the two. The transmission gain peak at M2 starts to shift down slightly remaining a relatively constant insertion loss of ~ 22dB before starting to fall-off at separation distances over 30cm. The S21 series and parallel modes start to move closer together with increasing coil situation, and remains in sync with the progressive narrowing of the parallel modes in the input impedance Z11 at M1 and M3. The phase response of the different resonant modes shifts accordingly and remains consistent with the gradual reduction in the induction field influence between the primary and secondary coils.
Overall when the sequence is observed it is clear to see that increasing the distance between the two coils is reducing the inter-action between the two, gradually separating them from a coupled coil system, to two independent coils with defined individual characteristics. It is interesting to note that the collapse of the coupled coil characteristics reflect changes that can be attributed to both the magnetic and dielectric induction fields. In the Z11 characteristics the two upper and lower parallel modes are gradually moving together in frequency, showing the reduced interaction between two modes at the same frequency, the upper from the primary at M3, and the lower from the secondary at M1. In accordance the series and parallel modes reflected in S21 are also proportionately moving together. The peak in S21 from the fundamental series resonant mode at M2, and the parallel mode at M4.
Figs 2.6-2.8. Show the final stages of collapse of the coupled coil characteristics as the distance between the two coils moves from 40cm to 60cm. Here the frequency axis has been zoomed to span only ~ 900kc, so that the details of the collapsing characteristics can be observed clearly. By 100cm the two coils are fully outside their field of influence, and the coupling of the induction fields between the two coils is insignificant, and the electrical properties of each are entirely dominated by the characteristics of the individual coil, and not by their inter-action. Any attempt to tune or adjust each individual circuit has no effect on the properties of the other. This may seem obvious since there is no-longer any coupling between the two coils, but the extent of the induction field influence is surprising at almost 1m between them, and suggests that the magnetic and dielectric fields of induction have a combined sphere of influence on the electronic properties of electrical elements, that can extend further than either of the induction fields individually.
Figures 3 below consider a simple equivalent circuit of the TC system modelled in LTSpice. The results of the modelling show the voltage transfer gain of the equivalent circuit over the frequency range 1 – 4 Mc. The modelled equivalent circuit reveals surprisingly close correspondence, for such a simple model, to the key features of both the series and parallel modes in the measured transmission gain S21, and especially using the actual measured and derived lumped element circuit values from the cylindrical TC.
The equivalent circuit consist of the following circuit elements:
L1 – The measured lumped element inductance of the secondary coil 350.7µH.
C1 – The total self-capacitance of the secondary coil derived from the fundamental series resonant mode at 2.27Mc, 14.0pF.
R1 – The series resistance of the secondary coil varied by the LTSpice model from 50Ω to 500Ω in steps of 50Ω to illustrate the effect of changing resistive losses on the transmission gain insertion loss, and quality factor Q of the resonant modes.
C2 – An element to represent the distributed inter-turn capacitance of the secondary coil, and including the conjectured extension of the dielectric field of induction across from the primary coil to the secondary coil. 12.2pF was required to model the parallel resonant mode to 3.33Mc, matching the measured parallel mode in the transmission gain S21 results.
R2 – The transformed up primary circuit resistance into the secondary, based on the TC turns ratio 24:3 and the measured magnetic coupling coefficient k ~ 0.27, R2 = 67Ω. This previously derived element value results in an insertion loss of ~ 5dB at the series resonant mode @ 2.27Mc. This matches very closely the insertion loss measured at this point in the S21 results.
Fig 3.1. Here the overall modelled characteristic can be easily recognised as most similar to the measured transmission gain S21 presented throughout this experimental post. The series resonant mode forms a transfer maximum at 2.27Mc and with an insertion loss ~ 5dB. The parallel resonant mode forms a transfer minimum at 3.33Mc and with an insertion loss ~ 73dB. The phase relation switches the model from inductive to capacitive at the series point, and then back to inductive again at the parallel mode. The phase relationship of the transfer gain moves through the complete ±90°. The variation of the series resistance of the secondary coil shows the changes in quality factor Q of the resonant circuit, and collapsing resonant modes with increased resistive losses. For such a simple equivalent model the match with the measured transmission gain S21 is good, and gives some insight into the nature and mechanisms of the Tesla coil under these conditions.
Fig 3.2. A zoomed view of the series resonant mode reaching a maximum at 2.27Mc, ~5dB insertion loss.
Fig 3.3. A zoomed view of the parallel resonant mode reaching a minimum at 3.33Mc, ~73dB insertion loss.
Fig 3.4. Here C2 has been removed, all other aspects of the equivalent circuit remain the same. This illustrates the effect on the transfer gain by removing the element for the distributed inter-turn capacitance, or that which is conjectured to form the parallel resonant mode, and which has most contribution to the formation of the LMD mode within the cavity of the secondary coil. The results show that the parallel mode is no-longer present, and this element is required to form the parallel mode characteristics in the coil. This suggests that the dielectric induction field is no-longer coupled across the windings of the coil, including across the windings from the primary coil to the secondary coil. The series mode resonance is not affected by this change showing how the parallel and series resonant modes, whilst stemming from the same coil geometry, have a relative degree of independence in the results, something that has also been noted in the experimental tuning and matching of the TEM and LMD modes for high-efficiency transference of electric power.
Fig 3.5. A zoomed view of the series resonant mode reaching a maximum at 2.27Mc, ~5dB insertion loss.
Overall the simple equivalent lumped element model shows interesting correspondence with the actual measured transmission gain, and helps to suggest and confirm the possible mechanisms involved in the formation of the series and parallel modes in a TC system. This model could obviously be developed to a much higher order, and it would be interesting to explore the modelled results for a complete TMT system, involving two matched resonant circuits, corresponding series and parallel mode splitting, and also the required elements necessary to represent the single wire transmission medium, if this is indeed possible in a linear Spice type model.
Summary of the results and conclusions so far
We have experimentally explored the transmission gain S21 for a cylindrical Tesla coil, compared and contrasted the results to Z11 (from S11) the input impedance of the TC, and found that the series and parallel resonant modes are both present within the system in both sets of measurements. A simple equivalent circuit model appears to support the understanding of how the series and parallel modes form, and their relative inter-action and inter-dependence or otherwise to each other. We have conjectured that the dielectric induction field is coupled across inter-turns of the primary and secondary coils, as well as between the primary and the secondary coil, and that indeed the complete picture of the Tesla coil requires both magnetic and dielectric induction to yield the fascinating and unusual phenomena demonstrated by TC and TMT systems.
When viewed as a whole system together both from S11 and S21 the TC is an induction transformer that extends both the magnetic and dielectric fields of induction from the primary to secondary. This is a most important point of consideration because it suggests that the very highest efficiency in the transference of electric power can be accomplished where the induction fields are in equilibrium and balance across the entire electrical system. If it is a TMT system that we are considering, then the highest efficiencies of transference take place when balance and equilibrium are established (tuned) for both the magnetic and dielectric fields of induction, extending all the way from the generator to the load, and both in the TEM mode in the two sections of the system, and in the LMD mode in the single wire and cavity sections of the system. The correct balancing and tuning of both modes allows maximum power to be transferred between source and load.
The analogy is to consider the TMT system as a tubular waveguide, or a pipe, between the source and load. In carefully balanced equilibrium the dielectric and magnetic fields of induction can propagate through the waveguide without experiencing discontinuous and abrupt changes in impedance of the waveguide, (the waveguide is not narrowing or widening along its length). In the LMD transmission model, the mode of transmission is being transformed from the TEM case to the LMD case, and where the waveguide transforms from a twin wire guide to a single wire guide. In the twin wire section the induction fields are in temporal phase but not spatial phase, where as in the single wire case the induction fields are in spatial phase, but not temporal. This phase temporal and spatial reversal and realignment between the mode transformations is for me the key to obtaining the highest transference of electric power in the TMT system.
Ultimately the case could be considered where the dielectric and magnetic fields of induction are coherent both spatially and temporally across the entire TMT system, from source to load through a balanced waveguide. This would lead to a coherent induction field condition where the magnetic and dielectric fields of induction are differentiated but coherent with each other, a condition for me that belongs to the principle of Displacement. Currently the most established macroscopic demonstration of this principle occurs in the field of superconductivity, where the magnetic and dielectric induction fields are differentiated but coherent across the material system, due to cooperation between the electronic and mechanical properties of the material. I conjecture that the inner workings of electricity are completely permeated with this coherent state of Displacement, both as a principle and a mechanism, of inclusive and coherent electric inter-action.
Of course this coherent state probably goes far beyond the basic electric properties of a system, but could be conjectured to be the next inner layer of the hidden, and underlying fabric of nature. Often referred to in the New Science or Alternative Energy fields as the “aether” or “aetheric field”, an amorphous energetic “field”, that is seemingly just outside material manifestation. It is claimed by some that this energetic field can be tapped through the correct principles and mechanisms applied to our experimental apparatus, and called-forth under specific conditions of coherence and particularly through non-linear events; the result of such conditions include, energy injection, and coherent phenomena that result in regenerative action, over-unity gain, and macroscopic coherence over vast spatial distance.
My own research work looks to progressively reveal the inner-workings of nature and these coherent phenomena, through exploring the principle of Displacement and Transference in electrical systems. This work proceeds through the inclusive union of high quality scientific experimentation, impeccable measurement, and considered conjecture in the outer world, and the inner quest for knowledge about my-Self, the hidden underlying wheelwork of nature, and our part within the great mystery of life.
1. A & P Electronic Media, AMInnovations by Adrian Marsh, 2019, EMediaPress
2. Dollard, E. and Energetic Forum Members, Energetic Forum, 2008 onwards.
In this second part on high efficiency transference of electric power, we take a look at the characteristics and power efficiency of a cylindrical coil TMT system where the transmitter and receiver coils are spaced further apart in the mid-field region. In this experiment a single wire transmission medium 11m long is used to separate the coils into different rooms at the laboratory, and a remote camera is used to observe the power at the receiver load measured by an RF wattmeter. Transference of electric power over 11m, and the characteristics of a TMT system coupled by the LMD mode at this distance, is shown to be remarkably different from the close mid-field region, and requires a very different setup and configuration of the experimental apparatus in order to optimise the efficiency of power transfer up to 96%.
In the close mid-field region with a 2m single-wire in the previous experiment on High-Efficiency Transference of Electric Power, the maximum transfer efficiency was achieved when the TMT system was configured, tuned, and operated at the point where the parallel modes were balanced, and the generator was optimally impedance matched to the system. It was conjectured that this balance contributes to maximising the power transferred from the generator to the twin-wire primary circuit TEM mode, to the single-wire LMD mode within the cavity formed between the transmitter and receiver secondary coils, and back to the twin-wire primary circuit TEM mode to the load.
In the mid-field region with an 11m single-wire we will see that this balanced mode setup leads to a maximum efficiency of ~40%. It is demonstrated that it is necessary to significantly mismatch the balance between the transmitter and receiver coils in order to get the LMD mode to extend across the single-wire transmission medium and restore transfer efficiency to over 90%. Transmitter and receiver primary circuit mismatch is mainly used to restore the transfer efficiency, along with fine adjustment through generator to TMT system TEM mismatch, measured at a range of Standing Wave Ratio (SWR) of 1, π/2, φ (the golden ratio), and 2.
The video experiment demonstrates and includes aspects of the following:
1. Small signal ac input impedance Z11 for a cylindrical coil TMT system in the mid-field region, and connected via an 11m 12AWG single wire transmission medium.
2. Z11 balanced parallel mode impedance measurements, for a reciprocal TMT configuration with 3 primary turns and matched primary capacitor tuning.
3. Z11 unbalanced parallel mode impedance measurements, for a non-reciprocal TMT configuration with 4 transmitter primary turns, 2 receiver primary turns, and mismatched capacitor tuning.
4. Transference of electric power from the linear amplifier generator to a 500W incandescent lamp load at the TMT receiver output via the reciprocal TMT configuration, and with a measured efficiency around 40%.
5. Transference of electric power to a 500W incandescent lamp load at the TMT receiver output via the non-reciprocal TMT configuration, and with a measured efficiency of up to 96%.
6. Demonstration of the high tension and associated discharge that can be drawn from the high-end of the receiver secondary coil, via the 11m single wire.
7. Transference of electric power efficiency measurements up to 96% (90% average) at 400W dissipated load power (peak 500W), in the 160m amateur radio band at 2.01Mc, and via an AWG12 single wire 11m long between the TX and RX coils.
Video Notes: The receiver power meter reading is shown on the inset video in the top right corner. For clear viewing and reading of the inset meter readings, and the VNWA software measurements, “720p” or “1080p” video quality is recommended, and may need to be selected manually from the settings icon once playback has started.
The experimental system circuit diagram, followed by an overview of the linear amplifier generator components is available here.
Figures 1 below show the key small signal input impedance characteristics Z11 presented in the video experiment, along with a more detailed analysis as to their impact on the observed and measured experimental results.
Fig 1.1. Shows the balanced and reciprocal input impedance for the cylindrical TMT system with 11m single wire transmission medium. The parallel modes, at markers M1, M2, M4, and M5, are balanced in the normal way by adjusting the primary tuning capacitors at both the transmitter and the receiver. The fundamental series resonant frequency M3 @ 2.02Mc has a series resistance RS = 11.3Ω, and is the primary drive point for the linear amplifier generator used in the experiment, with fine tuning around this point established at 2.01Mc as the optimum point. The parallel modes, one from the primary and one from the secondary, for both the transmitter and receiver coils are balanced, and show the frequency splitting that occurs when resonant modes of a very similar frequency are coupled together.
This form of impedance characteristic has been very well covered before in many posts on the website, and is discussed in detail in Cylindrical Coil Input Impedance – TC and TMT Z11. Previously these characteristics have been studied in the close mid-field region, typically with a single wire in the region of 1.5-2m long, or at least 2-3 times the diameter of the secondary coil, (0.5m in the case of the cylindrical TC). In this region the coupling between the transmitter and receiver coils, via the single wire transmission medium has been shown to be significant and the parallel modes split up to 200kc apart in frequency, as can be seen here. Within the split parallel regions there is a well defined and distinctive phase change from the extended series mode. The extended series modes, both upper and lower, can also be used as drive points for a linear amplifier generator, although the series resistance at these points is higher than the fundamental series mode, and ultimately will couple less total power from the generator through the TMT system.
With the single wire now extended to 11m in the mid-field region it can be clearly seen in this impedance scan that the coupling between the parallel modes of the transmitter and receiver has reduced, the frequency split is less at 30kc, and the extended series mode phase change is only just defined between markers M1-M2 and M4-M5. The fundamental series mode remains dominant at M3 and is the optimum drive point for linear amplifier generator. Overall the transmitter and receiver coils are coupled together by the single wire transmission medium in the TEM mode, but the coupling is reduced from the close mid-field region, and the additional impedance of the longer single wire is transformed back through into the transmitter primary and reflected in the increased series mode resistance at M3, RS = 11.3Ω.
Fig 1.2. Shows the effect of adding a 500W incandescent lamp load at the receiver primary coil output. The transmitter primary tuning capacitor CPTX has been adjusted from 663pF to 711pF in order to balance the transmitter parallel modes. The receiver primary tuning capacitor CPRX remains the same at 793pF. The resistive and inductive loading presented by the high-power incandescent lamp at the receiver has significantly changed the operating characteristics of the TMT system from a well balanced cavity, to a strongly unbalanced cavity, at least in terms of the TEM input impedance Z11.
The parallel modes of the receiver coil have been almost entirely suppressed with only a very slight presence at M3, and the overall resonant circuit properties of the receiver distorted and skewed away from the reciprocal coil characteristics of the unloaded receiver TC, to the characteristic shown at M3. It is important to note that this huge imbalance in the receiver end of the cavity in both the TEM mode, and I would conjecture the LMD mode due to the definite and distinctive change in the parallel modes, leads to a setup in this experiment where the transmitter end also needs to be unbalanced in order to reestablish the maximum efficiency in the transference of electric power. It is conjectured and discussed later that the setup change to the transmitter establishes a balance again in the LMD mode in the cavity when the total effect of the receiver and the longer single wire are taken into account together.
The fundamental series resonant mode has shifted down very slightly to 2.01Mc, RS = 13Ω, which was found to be the optimum drive point for the linear amplifier generator during the tuning and setup part of the experiment prior to the video experiment itself. The balanced reciprocal setup shown in figures 1.1 on this page, and 2.1 here , which was so effective in the close mid-field region, is shown to yield a maximum power transfer efficiency of now more than 35-45%. It is clear that the coupling introduced by the single-wire transmission medium and the impedance that this presents to both the TEM and LMD mode is critically important in both the setup and operation of a TMT system over distance.
Fig 1.3. Here the setup of the transmitter and receiver has been changed from that of the balanced reciprocal cavity condition, which yields power transfer efficiencies no higher than 35-45%, to the seemingly mismatched characteristic that yields measured transfer efficiencies up to 96% in the experiment. This setup requires the transmitter primary turns to be increased from 3 to 4, and a significant increase in the primary tuning capacitor CPTX = 1206pF. In correspondence, the setup of the receiver primary turns is also decreased from 3 to 2, and the primary tuning capacitor is significantly reduced to CPRX = 146pF. In this setup the input impedance Z11 for the TEM mode appears highly imbalanced, however for the LMD mode it is conjectured that a strong coupling and balance is re-established.
The fundamental series resonance at M3 has again only shifted very slightly in frequency to 2.0Mc, as the wire length of the experiment, the biggest contributor to this mode, remains constant, and with an increased series resistance RS = 22.8Ω. This still represents the best generator drive point for this experiment, with the lowest series resistance, and maximum coupling to the both the series and parallel modes that are active in this configuration. Transmitter parallel modes at M1, M2, and heavily suppressed around M3 and M4, are shifted quite considerably by the primary tuning capacitor mismatch. The dominant parallel modes, and hence conjectured to contribute most strongly to the LMD mode in the cavity, are now at M1 and M2 and involve both the transmitter and receiver, which will become apparent in the next figure. It should be noted that this figure is on a vertical magnitude of impedance scale of 4kΩ, whereas the previous figures where set to 1.5kΩ. This emphasises the very strong lower parallel modes and suggests that the transmitter pump action, from the generator to the LMD mode in the cavity, has been preferentially increased at this lower frequency of 1.2Mc.
The reduction in the primary setup at the receiver appears to have loosened the coupling between the primary and secondary coils of the receiver, which in turn has increased the Q of the free resonance in the secondary coil, increasing the phase change at M3, and emphasising the receiver characteristics transformed across the single wire cavity back to the transmitter. In short it appears like the LMD pump action into the cavity has been increased, whilst the Q of the receiver has also been increased. It is conjectured here that this combination of effects re-establish a balanced condition for the LMD mode, and hence a low impedance path for this mode across the cavity. With the LMD mode established across the cavity the efficiency of power transfer is pushed right back up to 95+%. Losses in the TEM mode are clearly increased with the longer single wire, but it is conjectured this is not the case for the LMD mode which is coherent spatially but not temporally over the entire cavity.
The split in frequency between the fundamental series mode at M3 and the upper extended series mode at M4 is now only 80kc, which is a very different condition than that which occurs in the balanced non-loaded mode. This close correspondence between these series two modes at the transmitter and receiver suggests part of the mechanism that allows very high-efficiency transference of electric power, where power is coupled from the primary to the secondary and hence into series modes to parallel modes, and then back through parallel modes to series modes at the receiver, a transformation across the TMT system from TEM to LMD and back to TEM mode in the load. Ultimately real power is passed from the generator through to the load which requires the TEM mode in both primary circuits, and the LMD mode as a result of the combined LM and LD modes across the cavity of the TMT.
Fig 1.4. Here we see a zoom of the peak of the dominant parallel mode from the previous figure at M1 and M2. Very interestingly we see that this peak is actually split into two peaks, suggesting two parallel modes that are dominant in both the transmitter and receiver but very weakly coupled. This now sets up the condition that we have two parallel modes separated by only ~ 1kc, and two series modes separated by only 80kc, from both the transmitter and receiver. I conjecture that it is this combination of series and parallel modes at each end of the TMT that makes it possible to yield very high-efficiency transference of electric power in this TMT system with a longer single-wire.
So what appears to be a loaded and unbalanced setup actually yields a TMT system that is balanced and matched for both the TEM and LMD modes combined. From a TEM perspective of the input impedance Z11 this appears to be heavily loaded and biased towards the transmitter, but on closer inspection and analysis suggests a configuration that balances the system between transmitter and receiver for maximum efficiency, minimum impedance for power transfer, and optimal conditions for the 500W incandescent load used in the experiment. Fine tuning of this configuration was further demonstrated by introducing a non-zero reflection coefficient from the transmitter primary circuit to the generator. This was accomplished by progressive adjustment of the antenna tuner away from the optimum SWR of 1.0, increasing up to 2.0. A standing wave ratio of π/2 to φ (the golden ratio) were found to increase the efficiency slightly making the difference between a stable 90% efficiency up to a maximum in this experiment of 96%.
It is suggested here that the TEM mismatch at the transmitter primary circuit is a method of fine tuning the balance of the circuit for the TEM and LMD modes combined. The balance between these two modes, and hence the energy coupled into and between these modes, and across the complete TMT system and cavity, appears to have the most impact on the power transfer efficiency.
Summary of the results and conclusions so far
In this post we have experimentally observed high-efficiency transference of electric power sustained at 90%, and with fine tuning and adjustment up to a maximum of 96% with an estimated error of ±1%. The power was transferred using a cylindrical coil based TMT system, where the transmitter and receiver are coupled by an 11m single wire transmission medium. 400W of power could be stably passed from the linear amplifier generator to the incandescent load at maximum transfer efficiency (90-96%), and up to 500W was tested at a reduced efficiency ~85%. From the experimental results and measurements presented the following observations, considerations and conjectures are made:
1. The “ideal” balanced reciprocal cavity setup, optimal in the close mid-field region, is not efficient for optimum power transfer in the more distant mid-field region, and most specifically when driving a heavy load at the receiver output.
2. An unbalanced TEM setup at the transmitter and receiver coil appears to restore the overall combined balance of the TEM and LMD modes across the entire TMT system restoring the high-efficiency power transfer characteristics in the mid-field region.
3. The unbalanced TEM setup appears to increase the LMD pump action into the cavity, whilst the Q of the receiver has also been increased by loosening the primary receiver coupling. It is conjectured here that this combination of effects re-establish a balanced condition for the LMD mode, and hence a low impedance path for this mode across the cavity.
4. The Z11 impedance characteristics in the unbalanced setup and when loaded at the receiver with a 500W incandescent lamp show a fine split between the series modes and the dominant lower parallel modes, which appears to show the transmitter and receiver coupled together in both the TEM and LMD modes
5. This close correspondence between these modes at the transmitter and receiver suggests part of the mechanism that allows very high-efficiency transference of electric power, where power is coupled from the primary to the secondary and hence into series modes to parallel modes, and then back through parallel modes to series modes at the receiver, a transformation across the TMT system from TEM to LMD and back to TEM mode at the load.
6. The maximum transfer efficiency could be fine tuned by mismatching the generator to the primary transmitter circuit and hence creating a reflection coefficient in the transmitter part of the system. SWRs in the region 1 to 2 were tested, with the best results around π/2 to φ (the golden ratio).
7. It is suggested, but needs considerable further work to develop, that the impedance presented by the single-wire transmission medium to the LMD mode is not the same as that presented to the TEM mode, and where a narrow single wire to the limit of the skin depth would appear as a high impedance at the driving frequency to the TEM modes, this is not the case for the LMD modes. For the LMD modes (LM and LD) the single-wire appears as a low impedance monopole waveguide which is spatially coherent over the extent of the cavity.
This experiment has opened up a range of interesting questions that need further consideration and considerable investigation to answer and progress, and most particularly from conclusion 7; to understand and establish in more detail the impedance presented by a single-wire transmission medium to the LMD mode generated in the cavity. It would also be interesting to compare the single-wire to a Telluric transmission medium, which will be the focus of the next experiment in this series. This experiment will look at transference of electric power over a 40m single-wire where the transmitter and receiver are in separate buildings of the lab, and also to compare the measured performance to a Telluric connection between the two via a basic ground system at each end.
1. Tesla, N., Colorado Springs Notes 1899-1900, Nikola Tesla Museum Beograd, 1978.
2. A & P Electronic Media, AMInnovations by Adrian Marsh, 2019, EMediaPress
3. Dollard, E. and Energetic Forum Members, Energetic Forum, 2008 onwards.