Multiwave Oscillator Impedance – Part 1

7th January 2019

The original Lahkovsky Multiwave Oscillator (MWO) apparatus combines two Tesla style drive coils in a transmitter and receiver configuration, each consisting of a primary and secondary coil cylindrically mounted on axis. The top-load for both transmitter and receiver is a complex combination of concentric half-wave resonators. The impedance characteristics of even a single drive coil with top-load represents a complex measurement challenge with results that can span over a very wide frequency range, in the order of 100kc/s – > 1Gc/s.

In this first part the small signal impedance characteristics, Z₁₁ (magnitude and phase) with frequency as seen by the generator, are measured for a single drive coil both with and without the MWO top-load over the lower frequency range of 100kc/s – 20Mc/s. In this measurement the impedance characteristics are dominated by the drive coil which will mask any higher frequency measurements pertaining to the MWO top-load. For this reason the top-load is measured individually in part 2 in the frequency range 100kc/s – 1.3Gc/s. Subsequent parts will look at the overall MWO system impedance characteristics when both transmitter and receiver are combined together in the original Lahkovsky arrangement, and later in an optimised and balanced drive arrangement as designed and presented by Dollard^[1].

In this first part the following measurements are presented:

1. Z₁₁ (magnitude and phase) with frequency for a drive coil without MWO top-load in the range 100kc/s – 20Mc/s

2. Z₁₁ for the drive coil combined with MWO top-load, and over the same frequency band.

3. Primary tuning measurements to match the resonant frequency of the primary coil, (with series loaded Cp), to the secondary coil at the fundamental, second, and third harmonics.

The SDR-Kits Vector Network Analyser 3E (VNA-SDR) was used to make all Z₁₁ measurements, and the apparatus and method of measurement is shown in Figures 1 below.

Fig. 1.1 Experimental setup for the MWO TC drive Z₁₁ impedance measurements, showing the TC drive coil, VNWA-SDR, calibration modules, and connection to the computer.
Fig. 1.2 The TC drive coil is connected via a pcb feed adapter to the VNWA-SDR via a short SMA RG178 coaxial cable.
Fig. 1.3 The TC drive adapter pcb is arranged to feed the TC via an SMA connector which allows for a calibrated series capacitance to be connected, and hence different resonant tuning points between the primary and the secondary.
Fig. 1.4 The feed pcb enables the VNWA-SDR calibration to extend right to the input terminals of the TC, eliminating spurious measurement effects from the connectors, cables, and geometry.
Fig. 1.5 The pcb feed adapter is calibrated in stages against a standard short circuit, open circuit, and 50Ω load. The series capacitance connector is here shorted with an SMA short when not being used.
Fig. 1.6 Here the feed pcb is connected to both the VNWA-SDR and a calibrated series capacitance box Cs. With both connected the VNWA was first calibrated using the three standard loads with Cs set to 1uF.
Fig. 1.7 The TC drive coil connected with MWO topload and measured in the same way as before to compare the impedance loading effects.
Fig. 1.8 The MWO topload is connected directly to the TC drive coil via short copper spacers where one end is driven from the secondary coil output, and the other open circuit, forming a λ/4 or shorter driven outer ring.

To view the large images in a new window whilst reading the explanations click on the figure numbers below.

Fig 1.1. Shows the overall measurement setup with the VNA-SDR connected to the drive coil by a short SMA terminated RG316 cable, and the other end to the coil feed adapter, and standard calibration modules.

Fig 1.2. The drive coil used in these measurements has 4.5 primary turns of AWG 10 ~2.5mm diameter magnet wire, and the secondary has 248 turns of AWG 24 ~0.51mm diameter magnet wire. The top-end of the primary coil is connected directly to the bottom-end of the secondary coil and forms the negative or ground terminal. The positive or drive terminal is connected to the bottom-end of the primary coil, and the top-end of the secondary can be seen emerging from the coil through a brass bolt which attaches to the driven top-load. Both coils are wound anti-clockwise on the former from the base.

Fig 1.3. In order to make an accurate measurement of Z₁₁ it is necessary to calibrate the VNA-SDR as close to the drive coil as possible. In this case the calibration plane is extended to the input of the primary coil terminals, (two 4mm high voltage shielded terminals), via a signal feed adapter pcb (SFA). The SFA can be removed from the drive coil by drawing the two-pronged 4mm probes out of the drive coil terminals. With the SFA disconnected the VNA-SDR can be calibrated by fitting an open, short and 50Ω standard load to the end of the SFA. The effective calibration plane then becomes the input to the drive coil, and spurious impedance effects due to any cables and the SFA itself can be removed from the final results. When calibrated, a frequency scan of the SFA with the 50Ω standard load will show a flat impedance line for |Z|, (magnitude of the impedance). The phase of this scan will swing repeatedly between ±180° indicating the near perfect match between the calibration plane and the 50Ω standard load.

Fig 1.4. Shows a close-up of the SFA connected directly to the two high voltage drive terminals in the base of the drive coil. The SFA has an SMA input feed and is then connected via equal weights of copper to the series connection point. The black terminal indicates the negative or ground point where both primary and secondary are connected together, and the red terminal the high voltage feed end of the primary. The series connection point in the positive terminal allows for a calibrated capacitance box to be connected in the primary circuit for tuning measurements. In this picture the series connection is not being used and is terminated with a SMA short. In tuning measurements, when series capacitance C_S is added, the SFA must first be calibrated with the capacitance box connected to the SFA with a nominal 1µF set at its output. The 1µF series capacitance has a very low impedance at the measurement band of interest and acts effectively as a short-circuit of the series connector during the calibration procedure. During tuning measurements the capacitance of the box is reduced in the range 100pF – ~ 50nF. The capacitance box itself uses surface mount standard capacitance values and can be reasonably used with SMA connection up to ~ 100Mc/s. The SFA is an unbalanced feed adapter and takes an unbalanced coaxial cable input directly to a balanced two terminal output without any compensation for this change in balance. A similar SFA was also tried which incorporated an RF (upto ~ 3Gc/s) balun in order to effect the transformation between the unbalanced and balanced connections. However, even with calibration the balun SFA proved to be less effective to measure Z₁₁ accurately and cleanly, as it dominated the impedance changes in the frequency band masking changes due to the drive coil further downstream. The standard SFA was therefore used with careful calibration up to the intended reference plane at the input terminals to the coil.

Fig 1.5. Shows the calibration procedure where the SFA is connected in turn to the standard calibration modules, here connected to the short circuit module. In this case the series capacitance terminal is not being used and is shunted with an SMA short.

Fig 1.6. Shows the SFA in measurement configuration and after calibration, where a 1nF series capacitance is added to the primary feed. The calibrated capacitance box can be adjusted in 1pF increments in the range ~30.5pF – ~10µF, where the parasitic capacitance of the box with SMA cable and when set to zero is ~30.5pF.

Fig 1.7. Shows the overall measurement setup with the top-load connected, and then calibrated and measured in the same procedure as previously discussed. The outer ring of the top-load forms an effective wire extension and capacitive load to the top of the secondary coil, which will both reduce considerably the fundamental resonant frequency and Q-factor of the drive coil. When compared with the drive coil without top-load the outer ring forms a λ/4 driven ring, which is coupling to 11 other λ/2 resonating rings. Given the size of the outer driven ring, and the proximity of the inner rings the coupling factor is considered to be quite high between the driven ring and the inner resonators, which will tend to constrain the free resonation of the inner rings and considerably quench the Q and hence measurements at much higher frequency. This is considered more closely in part 2.

Fig 1.8. Shows in more detail the mounting points at the top-end of the secondary and the MWO top-load.

Figures 2 below show the Z₁₁ impedance characteristics of the calibrated reference plane, the drive coil, and primary and secondary tuning with series added capacitance, of the drive coil only.

Fig. 2.1 Calibration results for the 50Ω standard load over the measurement window of 100kc/s - 20Mc/s. This calibration takes into account the TC feed pcb and a series capacitance Cs = 1uF, creating a calibrated reference plane at the TC input connectors.
Fig. 2.2 Magnitude and phase for Z11 showing the fundamental resonant frequency of the TC at M1 (1.28Mc/s), and many harmonics, (the VNWA-SDR software allows for a maximum of 12 simultaneous marker points).
Fig. 2.3 Narrowband calibration for Z11 against the 50Ω module over the range 100kc/s to 5Mc/s, to measure with greater detail the fundamental resonant frequency, 2nd, and 3rd harmonics.
Fig. 2.4 The fundamental resonant frequency, 2nd, and 3rd harmonic for the TC drive with a series primary capacitance Cs = 1uF, (very low impedance across the band).
Fig. 2.5 The series capacitance in the primary reduced to 400pF to show the 180° phase change in the primary that occurs at resonance (M4), and the accompanying very low series resistance at resonance.
Fig. 2.6 Cs has been increased to 540pF where the resonant frequency in the primary matches the resonant frequency of the 3rd harmonic of the secondary at M4 (4.31Mc/s). The matched primary and secondary resonance cause frequency splitting in Z11 at M3 and M5.
Fig. 2.7 Cs is further increased to 1200pF where the primary and secondary match now occurs at the 2nd harmonic M3 (2.95Mc/s).
Fig. 2.8 The primary and secondary resonance now match at the fundamental resonant frequency at M2 (1.28Mc/s), and Cs = 6500pF. When driven by a non-linear generator such as a spark gap the primary would resonate predominately at M1 (1.17Mc/s) with many other harmonics and beat frequencies present.

Fig 2.1. Shows the calibrated reference plane of the SFA when connected to the 50Ω standard load over the wideband range 100kc/s – 20Mc/s. The calibration is as expected and accurate over the band, with a slight phase variation between ~ 3 – 3.5Mc/s indicating a resonance in the signal path between the calibration plane and the VNA-SDR which has been normalised out by the calibration procedure.

Fig 2.2. The impedance characteristics of the drive coil with the series capacitance Cs = 1µF, showing a wealth of resonant peaks and corresponding phase changes from the fundamental resonant frequency F_S of the secondary coil at 1.28Mc/s up to the 12th harmonic F_S₁₂ (M₁₂) at 16.51Mc/s. It is interesting to note that the 2^nd harmonic F_S2 at M₂ is a stronger resonance than that of the fundamental at M₁. This is further demonstrated in subsequent impedance results. The VNA-SDR allows for only 12 concurrent markers which is why the final two harmonics on the results are not marked. At frequencies above 20Mc/s the rapidly increasing inductive impedance of the primary masks out any further harmonic points of interest, and the impedance curve rises rapidly until the primary coil becomes self-resonant at ~38Mc/s.

Fig 2.3. Shows the calibrated reference plane of the SFA when connected to the 50Ω standard load over the narrowband range 100kc/s – 5Mc/s. With more accurate calibration in the narrowband it can be noted that the phase variation seen in Fig. 2.1 has now been completely normalised out of the reference plane.

Fig 2.4. Shows the narrowband impedance characteristics for the fundamental F_S and the first two harmonics F_S2 and F_S3. Here the difference in magnitude of the resonance between F_S and F_S2 can be very clearly observed, although the Q of the coil remains very similar for both frequency points. The magnitude of the impedance |Z₁₁| ~ 28.5Ω at F_S, and ~ 49.7Ω at F_S2would significantly impede currents developed in the primary from the generator, coupling very little power to the secondary and ultimately to the MWO top-load. The primary circuit will need to be brought into resonance with the secondary at the corresponding frequency in order to considerably reduce the driving impedance seen by the generator, and hence maximise the primary currents, and the amount of power transferred from the generator through the drive coils to the top-load. On the very far left of the scan at 100kc/s the 180° phase change F_Ø180 of the primary in resonance with the series capacitance C_S can just be observed. This will be adjusted by varying C_S in subsequent results in order to tune the primary resonant frequency to that of the secondary. Given the difference in magnitude between the fundamental at F_S and the 2^nd harmonic at F_S2 it may be more optimal to arrange the primary to resonate at F_S2 rather than the fundamental for normal experimental operation. This will be tested during experiments and reported in subsequent parts.

Fig 2.5. Here the series capacitance C_S has been reduced from 1µF to 400pF which has increased F_Ø180 (the resonant frequency of the primary to M₄ at 4.93Mc/s. At this point the impedance seen by the generator has reduced drastically to ~0.72Ω allowing for much larger primary currents. In this case F_Ø180 does not correspond to a resonant frequency in the secondary so reduced power would be transferred between the two coils. At lower frequencies in the measurement band, the impedance of C_S is higher and has the effect of attenuating currents in the primary reducing considerably the magnitude of the lower resonant points, in this case considerably suppressing the fundamental resonance at F_S. This scan clearly shows the effect of including a series resistance in the primary and making it resonant within the measurement band of interest.

Fig 2.6. Here C_S has been increased to 540pF which has brought the resonant frequency of the primary equal to the 3^rd harmonic resonance of the secondary at M₄at 4.31Mc/s. As always with coupled resonant circuits, mixing of the signals causes beat frequencies and hence frequency splitting into two sideband frequencies where the impedance minimum points are at both the lower and upper resonant frequencies F_L3 (M₃) and F_U3 (M₅) respectively. If driven by a generator with C_S = 540pF F_L3 would be the best driving point since proximity coupling to the secondary top-load, or discharge of stored energy on the top-load, would momentarily increase the loading on the coil reducing slightly its secondary harmonic frequency towards F_L₃, and hence maximising transference of electric power between the primary and secondary coils.

Fig 2.7. Shows C_S increased further to 1200pF which now tunes the primary resonance to the 2^nd harmonic F_S2 at M₃ a frequency of 2.95Mc/s. Again frequency splitting occurs and the impedance of F_L2 and F_U2 are very low ~0.5 – 1Ω facilitating high current drive from the generator.

Fig 2.8. Shows C_S finally increased to 6500pF which tunes the primary to the fundamental resonant frequency of the secondary F_S at M₂ at 1.28Mc/s. At F_L (M₁) the lowest impedance drive point is obtained of just 0.34Ω at 1.17Mc/s. Given the very similar minimum |Z| characteristics of both F_S and F_S2, when tuned respectively to the primary with corresponding values of C_S, either the fundamental or 2^nd harmonic could be used to operate the coil in experiments to explore MWO operation and effects.

Figures 3 below show the Z₁₁ impedance characteristics of the calibrated reference plane, the drive coil, and the primary and secondary tuning with series added capacitance, of the drive coil with the MWO top-load.

Fig. 3.1 Same wideband Z11 impedance characteristics for the TC drive now including the MWO topload. The fundamental M1 has moved down significantly from 1.28Mc/s to 650kc/s and is also quite strongly suppressed. The 2nd harmonic has strengthened, and the other higher harmonics remain mostly unchanged.
Fig. 3.2 Narrowband detailed impedance scan for the fundamental resonant frequency, 2nd, and 3rd harmonics.
Fig. 3.3 Cs = 400pF which represents a higher impedance at lower frequencies, and almost entirely suppressing the fundamental resonance.
Fig. 3.4 Cs = 570pF a slight increase, (previously 540pF without the MWO), in series capacitance in the primary in order to match the primary resonance with the secondary 3rd harmonic.
Fig. 3.5 Cs = 1300pF again a slight increase, (previously 1200pF without the MWO), in series capacitance in the primary in order to match the primary resonance with the secondary 2nd harmonic.
Fig. 3.6 Cs = 26000pF a very large increase, (previously 6500pF without the MWO), in series capacitance in the primary in order to match the primary resonance with the secondary fundamental.

Fig 3.1. Shows the wideband scan for the drive coil with MWO top-load on the same horizontal and vertical scaling as in Figures 2. The top-load at lower frequencies represents a considerable capacitive load at the top of the coil, very similar capacitively to a toroidal, cylindrical, spherical, or sheet metal top-load added to a conventional Tesla coil apparatus. The frequency of this scan is too low to show any of the high frequency features that result from the resonant rings of the MWO, (see part 2 for these features), but the capacitive loading of the MWO top-load could be clearly seen in this lower frequency scan. The biggest effect is noted at M₁ at F_S1 the fundamental resonance of the secondary which has reduced from 1.28Mc/s to 650kc/s, a 49.2% reduction . The 2^ndand 3^rd harmonics at F_S2 and F_S3 respectively have also reduced but in a much smaller range than the fundamental. F_S2 has reduced from 2.96Mc/s to 2.84Mc/s a 4.1% reduction, and F_S3 from 4.32Mc/s to 4.21Mc/s a 2.5% reduction. Higher harmonics also reduce with progressively reducing amounts up to F_S12 at M₁₂ which reduces from 16.51Mc/s to 16.46Mc/s a 0.3% reduction. It can be seen for this arrangement of drive coil the fundamental resonant frequency would be very sensitive and easily effected by metal loading to the top end of the secondary, the close proximity of metallic structures, or even to the proximity of partially conducting mediums such as the human body.

Fig 3.2. The narrowband scan shows the large change in the fundamental in relation to the 2^ndharmonic both in frequency shift and in the magnitude of the impedance and phase change. The MWO top-load has considerably quenched the fundamental of the drive coil, whilst leaving the 2^nd harmonic very similar to the coil only results (Figures 2) both in frequency and the magnitude of the impedance and phase.

Fig 3.3. Series capacitance C_S = 400pF as before shows the primary resonance in the band, but in this case has completely quenched the already weak fundamental resonance. It can be seen that driving the MWO at the 2^nd harmonic during experiments and operation may lead to considerably more stable and reliable system characteristics.

Fig 3.4. Shows the primary tuned at F_S3 at M₄ at 4.20Mc/s and a slightly increased series capacitance required to tune, increased from 540pF to 570pF.

Fig 3.5. Shows the primary tuned at F_S2 at M₃ at 2.83Mc/s and again a slightly increased series capacitance required to tune, increased from 1200pF to 1300pF. It can be noted that the fundamental resonance F_S can now just be discerned at M₁ at 650kc/s.

Fig 3.6. Shows the primary tuned at F_S at M₂ at 0.65Mc/s with F_L = 610kc/s and F_U= 700kc/s. The primary is tuned when C_S = 26000pF (26nF) which is a very large increase on the 6500pF required without the MWO top-load. At F_L(M₁) the impedance seen by the generator in the primary is now 0.61Ω which is no-longer the lowest driving impedance. With C_S = 1300pF and tuned to F_S2 the driving impedance at F_L2 is 0.47Ω which indicates that better power transfer from the generator to the secondary could be effected by driving the MWO at the 2^nd harmonic. At this point the system would also be more stable and less effected by proximity of metal structures and other partially conductive mediums including the human body.

The lower frequency impedance measurements have shown a wealth of frequency harmonics associated with the Tesla style drive coil, and can be tuned progressively with the primary of the drive coil via series capacitance to make use of the system at different designed resonant frequency points. The MWO top-load at lower frequencies introduces a considerable capacitive load on the drive coil which has a large quenching effect on the fundamental resonant frequency of the system, and suggests a shift of optimal setup away from being driven at the fundamental and towards the 2^nd harmonic where greater transference of electric power should be possible, and the system would be more stable and tolerant of loading conditions on the drive coils, the MWO top-load directly, and also from proximity of other conducting and partially conducting mediums. It will be interesting to determine if this changes when both the transmitter and receiver are combined in the full Lahkovsky MWO system, and also whether these effects can be directly demonstrated through experiments in the operation of the complete MWO system.

Click here to continue to part 2 of the multiwave oscillator impedance measurements.

1. Dollard, E., Design and presentation of an optimised and balanced MWO power supply and drive coils., Energy, Science, and Technology Conference (ESTC), 2018.

2. Vril Science, Lahkovsky Multiwave Oscillator, 2019, Vril

Generators, Vacuum Tube

Vacuum Tube Generator (811A)

1st January 2019

The vacuum tube generator (VTG) mainly used for experiments in the displacement and transference of electric power is based around a pair of 811A power triode vacuum tubes, of either RCA or Russian origin, and with electrical characteristics generally as defined in the RCA 811A datasheet. The 811A’s have demonstrated to be highly flexible, with high reliability, and good overall medium power performance, versus cost and availability, when used in a variety of different configurations. The final output of this generator with these tubes can provide a maximum sustained RF output power of ~600W, and peak output for short bursts (up to 10s) of 900W, and over a wide frequency band up to ~5Mc/s. In addition, the design and implementation of this generator has been arranged in such a way that it can be used in a variety of different configurations, including:

1. A tuned plate class-C Armstrong oscillator which derives automatic feedback from a pick-up coil placed close to the secondary coil.

2. A variable frequency Hartley power oscillator when combined with an independent oscillator drive module.

3. The output stage of a linear power amplifier, when fed with a suitable drive waveform in grounded grid, grid biased, or cathode follower configurations.

4. A modulator stage with suitable cathode keying via a mechanical or semiconductor switching circuit.

5. CW, burst, and modulated oscillation modes when used with or without a dc smoothing capacitor in the plate HV supply.

Other VTGs using the RCA 833A/C and Eimac 4-400A/C are more suited to higher power experiments including, telluric transmission and strong non-linear impulses for displacement experiments, and will be reported in subsequent posts.

Figures 2 below shows an overview of the VTG apparatus, complete with the high voltage supply, the independent Hartley power oscillator module (HPO), the high voltage bridge rectifier (HVBR), and the supply dc smoothing capacitor.

Fig. 1.1 HV supply with the vacuum tube generator connected on the upper platform, and including HV rectifier on the left, and vacuum tube unit on the right.
Fig. 1.2 The VTG uses a pair of 811A tubes, is fan cooled, and can be configured as a power amplifier, oscillator, and rectifier, and in CW, burst, and modulated modes.
Fig. 1.3 HV supply (lower level) provides mains ac to the VTG to power the filament transformer, and 15V dc to power the fan.
Fig. 1.4 The VTG driven by a high power Hartley oscillator, making a very flexible variable band and frequency oscillator up to ~600W continuous, and ~900W peak power for short bursts.
Fig. 1.5 The Hartley oscillator drive unit showing the configurable HV fixed capacitors to set frequency band, and the HV variable capacitor to set frequency in the band. The oscillator can be tuned and drive the vacuum tubes over the range 1000-3700kc/s.
Fig. 1.6 The VTG connected to the high voltage bridge rectifier, and dc smoothing capacitor in the background.
Fig. 1.7 The input of the high voltage bridge rectifier is connected directly to the output of a single MOT, and its output connected directly to the dc smoothing capacitor.
Fig. 1.8 The dc smoothing capacitor is a Russian K75-40a 60µF 4kV pulse discharge unit.

The circuit diagram for the VTG and peripherals is shown in Figure 2 below, or click here to view the high-resolution version.

Fig. 2.1 Vacuum tube generator schematic showing the main generator, high voltage bridge rectifier, and the Hartley power oscillator module.

The high voltage supply V_HT (described here), is connected to the input of the HVBR. The rectifier consists of 4 sets of 2 x HVP20 20kV 750mA high voltage diodes in parallel and arranged in a bridge configuration. The output terminal of the bridge B is connected to a tank capacitor 60µF 4kV k75-40a Russian pulse capacitor which results in a smoothed dc output suitable for rapid charging and discharging through a primary, or as a stabilised dc supply for the VTG plate circuit.

A dc meter is also so arranged on the output B to read from 0-4kV in configurable steps based on where the base terminal of the meter is connected to a resistor divider. The high voltage dc meter is a simple arrangement using a 1mA fsd (full-scale deflection) ammeter connected to a resistor divider with 4 x 1MΩ resistors. With all four resistors in series 4kV will result in a 1mA passing through the meter moving it to fsd, with one resistor in series 1kV will result in meter fsd, and accordingly for the 2 and 3kV scales. The meter face has been recalibrated to indicate needle reading in kV.

Before considering each of the configurations listed above in detail, it is necessary to cover the basics of the VTG design. The plate supply is provided by the HVBR output B+ and is a variable 0-4kV peak supply either smoothed or unsmoothed based on whether the rectifier tank capacitor is connected or not. The plate supply B+ is connected directly to the driven load L_PC_P, which in this case is the primary coil in parallel with the primary tuning capacitor. In this arrangement the primary load (parallel resonant circuit) would present the highest impedance to the vacuum tubes when at resonance. When the primary capacitor is so adjusted to set the secondary resonance frequency it is optimal for the impedance of the primary load to be equal to the output impedance of the dual 811A vacuum tubes in parallel. In this way maximum power is developed in the primary load or through the primary of the coil.

It is important to note that when using a VTG the primary circuit, (in this case the load L_PC_P), is NOT arranged to resonate at the same frequency as the secondary circuit, which avoids large primary currents being developed in the VTG which would lead to excess plate dissipation in the vacuum tubes, and rapid degrading or destruction of the tubes. Rather the impedance of the primary load at the resonant frequency of the secondary coil defines the driving impedance presented to the vacuum tubes, which in turn are adjusted by the grid bias or feedback to match their internal impedance to the driving impedance, leading to maximum power being developed in the primary circuit whilst not overloading the vacuum tubes beyond their maximum ratings.

Hence the VTG drives the coil in a linear sinusoidal CW mode with optimally arranged load impedance matching, as compared with for example, a spark gap generator where the primary and secondary resonance frequencies are arranged to be equal to allow maximum power transfer from the maximised primary currents during the ring-down phase of the primary tank discharge.

The load of the primary is connected to the plates of the vacuum tubes via high frequency chokes L1R1 and L2R2. These chokes quench very high frequency oscillations which can be generated by discharges within the vacuum tubes during overload conditions, and in turn help prevent very high frequency oscillation run-away conditions which can lead to rapid tube destruction.

The cathodes of the vacuum tubes are bridged by RF bypass capacitors to minimise the impedance of the high frequency signal path, and then connected via a sequence of jumpers J1-J3 to allow for different configuration modes. The cathodes are further connected to the heater power supply which provides a constant bias to the cathode combined heater element of the 811A. The heater power supply is a mains step-down transformer so arranged to provide an ac supply of 6.3V_RMS @ 8A_RMS for two 811A heater circuits in parallel. The heater supply transformer is fine adjusted in this case by R5 which reduces the voltage across the heater elements and has an RF bypass capacitor C5 to again minimise the impedance of the high frequency signal path. The ac voltage across the tube heater elements is monitored using a calibrated 0-10V_RMSac voltmeter, and R5 adjusted, (in this case 4 x 0R1 25W series connected resistors), to provide an optimal 6.3V_RMS +- 5%.

When the vacuum tube generator is used as a linear power amplifier in grounded grid (cathode driven) configuration it is necessary to prevent the RF input signal from feeding back into the heater supply and being dissipated in the low impedance power supply stage. To prevent this bifilar high frequency chokes can be used between the tube cathodes and the heater power supply. At low-frequency, (50Hz for the heater supply), current can pass through the choke from the heater supply to the tube cathodes, but the RF signal at the cathode is prevented from feeding back into the heater supply. Modulation and switching via cathode keying can also be arranged via mechanical or semiconductor switches, and allows for a range of “switched” experiments important in the comparisons of electrical phenomena experienced in the exploration of the displacement and transference of electric power.

The grid of the VTG is arranged with jumpers J4-J6 to allow for different configuration modes, and a current limit resistor R3 to restrict the maximum grid current below the nominally rated for the 811A, and according to the configuration it is used in. In addition both 811A’s are forced cooled by the fan driven by the auxiliary 15V supply. Force cooling allows for higher sustained power in CW modes, and affords an additional protection during peak power overloads. It is quite normal for the 811A plate to glow slightly red under higher sustained powers, and in peak power for short periods 0-10s for it to glow intensely red. Sustained high peak powers > ~650W will lead to plate dissipation overload combined with flash-overs between the plate and grid causing rapid grid damage. In most experiments in the displacement and transference of electric power I have found a comfortable sustained power between 400-600W, which leads to long vacuum tube life, and well sustained tube characteristics according to the nominal data.

Construction of the VTG and HVBR modules are shown in detail in Figures 3 below. For ease and simplicity of experimentation the VTG is open assembled on a simple board with simple insulated mountings and a combination of metal, plastic, and wooden mounts for the various components. Whilst this does lead to a very quick and flexibly modifiable prototype generator, there are significant EMC, interference, temperature, and stability benefits to housing the entire VTG build within a screened metal case, with directed cooling inlet and outlets, and with careful consideration to connection of high current and tension paths with minimal inductance copper bars etc.

Fig. 3.1 VTG showing twin 811A tubes with fan cooling, filament transformer and ac filament voltage meter, grid bias components, and external connectors.
Fig. 3.2 Overhead view of the VTG showing layout of the various key components.
Fig. 3.3 Right side view of the VTG showing grid bias capacitors and power resistor, filament bias resistors, and cathode outlets.
Fig. 3.4 Rear view of the VTG showing plate and grid connectors, filament transformer, and filament high frequency chokes.
Fig. 3.5 Left side of the VTG showing the heater element transformer, meters, and the optional bifilar chokes.
Fig. 3.6 Overhead view of the high voltage bridge rectifier showing HV diodes, input and output connections, and adjustable kV output meter.
Fig. 3.7 Left side of the HVBR showing the inputs and outputs.

Figures 4 show the construction of the Hartley power oscillator module which is considered further in configuration option 2 below.

Fig. 4.1 High power Hartley oscillator module showing tuning capacitor, frequency band capacitors, and tapped inductor.
Fig. 4.2 Overhead view showing the layout and interconnection of components.
Fig. 4.3 Oscillator connections to VTG for plate supply (red), grid bias circuit (black), and rf ground (blue).
Fig. 4.4 Rear view showing the grid bias capacitor connected to the grid bias variable resistor.
Fig. 4.5 Lowest frequency band 1.5Mc/s - 1.8Mc/s with all three band capacitors connected, (2 x 1000pF, 1 x 500pF).
Fig. 4.6 Mid frequency band 1.8Mc/s - 2.4Mc/s capacitors connected, (1 x 1000pF, 1 x 500pF).
Fig. 4.7 Highest frequency band 2.4Mc/s - 4.1Mc/s with a single band capacitor connected, (1 x 500pF).

The different configurations of the VTG module are now considered in more detail:

1. Tuned plate class-C Armstrong oscillator

This configuration is very well suited to investigations at the upper and lower resonant frequency of the coil being driven (F_Uand F_L). In this case the VTG is a series-fed version of the Armstrong oscillator deriving the grid bias feedback from a pickup coil placed in proximity to the secondary coil. In the case of the flat coil this is a small 10 turn cylindrical coil (diameter 100mm) mounted behind the secondary coil and on axis with the coil centres. Oscillation via the pickup coil feedback automatically keeps the VTG on the resonant frequency being explored and adjusts automatically to maintain the resonant frequency when loading is applied to the secondary coil outputs.

In this configuration J1-J3 are left open and J4 is connected. Circuit operation is as follows. When first turned on the impulse current from the tank circuit B+ conducting through the vacuum tube causes a ringing oscillation (ping) in the primary circuit L_pC_p. This oscillation couples to the secondary circuit L_sC_s which is further coupled by the pickup coil to the grid bias leakage circuit. When the phasing of the pickup coil is the correct way round for positive feedback, the grid bias leakage capacitor C1 becomes negatively charged during the positive half cycle in the primary circuit, pushing the grid voltage down and progressively restricting conduction in the vacuum tubes towards the off state with much reduced plate current. The negatively charged grid leakage capacitor C1 then discharges through the grid leakage resistor R4. As this happens the grid voltage on the vacuum tube starts to rise progressively towards 0 volts turning on the vacuum tubes with an increasing plate current. The plate current through the primary circuit L_pC_p again is coupled to the secondary L_sC_s and the cycle repeats. With the grid bias leakage circuit correctly adjusted the VTG will oscillate with a linear sinusoidal output optimised for maximum plate voltage and current swing, (maximum power transfer at the resonant frequency of the secondary), whilst keeping the grid bias currents within the maximum ratings for the vacuum tubes used.

When setting up this mode of operation it is most important to ensure correct phasing of the pickup coil in connection to the grid bias leakage circuit, and that the values of C1 and R4 are suitably adjusted to allow maximum swing of the plate circuit whilst keeping the grid bias within the maximum ratings. For this VTG with 2 x 811A vacuum tubes R4 is optimally between 1-1.5kΩ and C1 between 1.5-3nF. Considerable power dissipation occurs in R4 when running the VTG at higher powers > 400W, hence the need for a wire-wound power resistor (100W), and preferably as part of the forced cooled air circulation. Excessive or too little grid bias current will lead to a distorted and clipped oscillation, or in extreme cases, no oscillation at all and rapid vacuum tube degradation or destruction. Initial setting up is best done with low plate voltage ~ 500V and higher values of R4. R4 can then be progressively reduced to increase grid bias whilst ensuring a clean and stable output oscillation.

As seen in previous impedance measurement posts for the flat coil (1S-3P) the resonant frequency of the secondary can be adjusted by changing the capacitance of the variable vacuum capacitor in the primary C_p. When adjusted the oscillating frequency of the VTG automatically changes to track the changes in the secondary resonant frequency. At high values of C_p > ~650pF the impedance of the lower resonant frequency F_L is dominant and the final oscillating frequency can be adjusted around this frequency F_L in the range 1.5-2.2Mc/s. At low values of Cp < ~650pF the impedance of the upper resonant frequency F_U dominates and the final oscillating frequency is in the range 2.7-3.8Mc/s.

Adjusted in this fashion this configuration of the VTG is very well suited to continuous linear measurements for single wire currents, displacement and transference of electric power, and telluric transmission experiments. This configuration is best suited to exploring frequency regions centered around F_U and F_L, where oscillation is stable and conduction currents in the secondary of the experimental coil are > 10mA_RMS. This configuration is not suitable for exploring frequency regions far from resonance, at very low bias currents, or between the transition between F_U and F_L. In these cases it is necessary to use the VTG in modes 2 or 3 where accurate progressive frequency control is provided by an external source and the VTG acts as a power stage, whether that be as a tuned power oscillator, or as a linear power amplifier.

2. Variable frequency Hartley power oscillator

The Hartley power oscillator converts the VTG to a linear oscillator which can be adjusted for variable frequency in bands defined by the combination of band capacitors connected on the HPO module. The frequency of oscillation of the VTG is now determined by the resonant circuit formed on the HPO board, the pickup coil of configuration 1 is not connected in this arrangement. This configuration is suitable for measurements across the entire frequency band of interest, in the case of the flat coil 1S-3P between 1.5-3.5Mc/s.

The HPO must be manually tuned or retuned to a specific frequency of interest and particularly when changing the loading of the secondary coil circuit. The resonant frequency of the secondary coil is very sensitive to electrical loading, temperature, material losses, changing boundary conditions and proximity, which all cause deviations of the configured frequency. Any change in this tuned frequency requires re-adjustment making the HPO configuration not well suited to experiments designed to explore different loads and operating conditions. It is however very suited to exploring circuit operation where the loading conditions are relatively fixed, and particularly in off-resonance, low current, and frequency transition regions.

In this configuration J1-J3 are left open and J5 is connected to the Hartley oscillator module along with the plate voltage and RF ground as shown on the circuit schematic of Fig 2. The bands of frequency can be adjusted by hardwired jumpers on the HPO module, where the static band capacitors are combined with the 1000pF variable capacitor, and form a parallel resonant circuit with L5 the HPO coil with an inductance of 3.0uH. The available bands are broadly as follows (static combinations of capacitors shown only):

a. 1 x 500pF = 2.4 – 4.1Mc/s

b. 1 x 1000pF = 2.1 – 2.9Mc/s

c. 1 x 500pF + 1 x 1000pF = 1.8 – 2.4Mc/s

d. 2 x 1000pF = 1.7 – 2.1Mc/s

e. 1 x 500pF + 2 x 1000pF = 1.5 – 1.8Mc/s

The setup of the HPO again requires a balance between largest plate swing (output power) without distorting the output wave, whilst restricting the grid bias within maximum parameters. The required frequency band is first configured using the hardwired jumpers, and the HPO grid bias variable resistor is set in the higher halve of its resistance range . The VTG plate supply is first set low at 500V and the HPO coil tapping point starts close to the RF ground end. The tapping point can be progressively moved upwards towards the plate voltage end of the coil until a point is reached where the output of the VTG is stable, clean, and with good power output as the plate supply is progressively increased up to the maximum ratings for the vacuum tubes. The HPO grid bias resistor can then be progressively reduced keeping the grid bias current within the maximum recommended. If no oscillation can be obtained the grid bias resistor can be progressively reduced until oscillation starts, whereupon the other adjustments discussed can be continued with.

The HPO can then be varied across its band frequency range according to the experimental requirements of the circuit, and the output power adjusted using the plate supply voltage. If the loading on the secondary coil changes significantly the HPO will need to be re-tuned and/or re-adjusted to provide stable power oscillation. Off resonance of the secondary coil in the experiment may require the HPO tapping point to be re-adjusted towards the RF ground to prevent excessive power dissipation in the coil. During continuous operation the temperature of the HPO inductor coil should be monitored in order to prevent over-heating. Re-adjustment will be required when moving often and rapidly between on-resonance and off-resonance operating points. As previously stated this configuration mode is best suited to exploration of operating points not accessible with configuration 1, and where the operating conditions do not change rapidly during the measurement cycle.

3. Linear amplifier output stage

The VTG can be operated as the power stage of a linear amplifier when correctly configured and connected to a suitable frequency generator with power amplifier output stage. No pre-amplifier stage is currently provided in the VTG so the driving signal source will need to generate an output between ~ 1-10W in order to the drive the VTG output to a usable output power. In this configuration the VTG can be driven in grounded grid, grid biased, or even as a cathode follower with some change of circuit connection and setup.

For grounded grid operation J1 is connected to the external signal source, and J6 is connected directly to RF ground. For grid driven mode J1-J3 are open and J5 is connected to the external signal source which in this case also needs to arrange for the driving signal to provide suitable dc grid biasing for the vacuum tubes used. For this reason grounded grid operation is preferred for simple linear amplifier operation.

When correctly setup and driven the VTG in this configuration can provide a very stable oscillation output which can be easily and finely adjusted by the external signal source, overcoming some of the adjustment problems of the HPO, but not exceeding the very good power output levels of the HPO. It has been found that the HPO is better when higher power experimentation is required, but the linear amplifier is better for overall signal stability and accurate adjustment. This configuration also allows for non-sinusoidal waveforms to be applied to the experimental system.

4. Modulator stage

Modulation of the VTG is currently by cathode keying, allowing conduction through the vacuum tubes to be switched at low frequencies via mechanical switches and relays, or at higher frequencies < 100kc/s by semiconductor MOSFET switches. Modulation of this kind was originally considered to be important in the exploration of the displacement of electric power where non-linear events play a very significant part in the unusual electrical phenomena observed within an electrical system. It has since been found through experimentation that the non-linear impulses generated by this modulation method are not of sufficiently low transition time and low pulse width to be particularly useful in the generation, observation, and measurement of displacement phenomena.

5. CW and burst oscillation modes

The VTG can be arranged in any of the configuration modes 1-4 and then operated in CW or burst modes simply by removing the plate supply tank capacitor at B+. When the tank capacitor is connected at the output of the HVBR a constant plate supply voltage B+ is applied to the primary load. In this case the output of the VTG in oscillation will be in CW mode providing a constant and continuous oscillation wave to the primary circuit L_pC_p.

When the tank capacitor is removed from the HVBR the output is an unsmoothed full wave rectified supply at 100c/s (based on UK line frequency). Applied to the VTG this produces bursts of oscillation inside the supply envelope, and has proven to be useful in establishing certain operating conditions in experiments orientated to displacement of electric power. These operating conditions and experiments are currently work in progress and will be reported in subsequent posts.

Overall the VTG has proven itself to be a versatile and reliable linear power source suitable to drive a wide range of experiments to explore the displacement and transference of electric power, and also some preliminary lower power telluric transmission experiments. Over several years of operation only 1 x 811A has been replaced after grid flash over destroyed one of the tubes when being used in configuration 2 with the HPO at an off-resonance operating point, with large output mis-match and high reflected power. The VTG has also sustained reliably through high power experiments where both plates of the tubes are glowing bright red for short periods of time. The 811A in RCA and Russian forms have proven themselves to be robust and reliable provided the heater element (spring tensioned) has not broken during extended storage. VTGs with other vacuum tube types and configurations will be presented in subsequent posts.

Click here to continue to the next part, looking at Tube Power Supply – Heater, Grid & Screen.

Displacement & Transference, Theories

Displacement and Transference

26th October 2018

As an experimental researcher it is normally always my preferred choice to share and discuss any theory I may hold about my work and the larger subject area according to the progression of the experimental work, and whether it corroborates or refutes any specific theory, principle, conjecture, or hypothesis I may hold. The principles that appear clear in my mind, regarding the displacement and transference of electric power, have guided the entire direction of my research efforts over a good many years to establish the validity or otherwise of these principles. In other words, I am designing and building the nature of my experiments in such a way as to attempt to reveal and test these working hypotheses and conjectures, and in so doing uncover and make further known the inner workings of the electrical wheel of nature.

Following interest and recent questions with regard to the nature of Displacement and Transference of electric power, the use of this terminology needs to be clarified in more detail, and ahead of the necessary supporting experimental results, which is work in progress at this time. The implications of these two mechanisms (displacement and transference) are vast, and part 1 of this topic is intended only as a summary and clarification of these principles as I see them, and to pave the way for more detailed discussion in subsequent parts, and of course further development as experimental results dictate.

It is important to first establish that with regard to displacement I am not referring to Maxwell’s displacement current, but rather to a more underlying phenomenon that precedes what we currently measure electrically via voltages and currents, and that which precedes the linear inter-action between the electric and magnetic fields of induction, or in other words the mechanism of transference.

In explaining what I mean by this I would first like to stress that these are working theories, hypotheses, and conjectures which are guiding me in a programme of experimentation to ascertain their validity or otherwise, and I am by no means claiming them to be true, tested, or proven. Experiments and results will establish or refute the validity of these theories all in good time. And even more, it is not enough for my own experiments to show the validity of these theories, but also they require experimentation and corroboration from others. How I have come to them is not easy to explain, other than to say that during many years of working with science and engineering, along with certain other subjects, and by studying electrical “over-unity” examples, circuits, and phenomena, they have come to me in the form of intuitive insights, light bulb moments, and after long nights trying to solve seemingly unrelated problems both theoretically and experimentally.

It is easier to discuss transference first as this can be readily measured, experimented, and understood from the huge edifice of knowledge available in the fields of electromagnetism and electrical and electronic engineering. In very short summary, transference refers to the electrical phenomenon that results from the linear inter-action of the electric and magnetic fields of induction, at best, spatially out of phase and temporally in phase, but overall an incoherent phenomenon.

This inter-action between these two fields is linear and whose results are understood very well by employing Maxwell’s four primary equations as synthesised by Heaviside^[1]. In turn this yields the telegrapher’s equation, the resolution of two linear differential equations, which can be used to very well model, simulate, and measure the electrical properties of a circuit network, and has been very well discussed and explored by Dollard^[2,3] and later EF^[4] in the form of the Heaviside equation:

(1) $\begin{equation*} ZY = h(RG + XB) + j(XG - RB) \end{equation*}$

In other words, electrical energy is transferred in a linear fashion (propagates) from one point to another in a well-defined time, and with well-defined characteristics, which result from the inter-action of the electric and magnetic fields of induction with the surrounding medium, materials, and boundary conditions. Transference is the common mechanism which yields the known and observed electromagnetic and electrical circuit properties, irrespective of the model by which the transference is accounted for, whether it be classical mechanics and electromagnetism, quantum electrodynamics, or other modern physical theory.

Transference will always result in discharge, dissipation, and ultimately loss of the available electrical energy to the surrounding and intervening medium (of which the circuit also belongs). Transference is the most basic mechanism by which electrical energy can be transferred from source to load in meeting the designed and prescribed purpose and “need” of the circuit. It is an incoherent mechanism, which always results in loss and at best a temporary rejuvenation of the system, and yet is currently our most “advanced” mechanism by which we can utilise electrical energy to do our work. Transference can be entirely measured via voltage and currents distributed over n different frequencies with n different phase relationships, and hence is entirely measurable with electrical and electronic equipment of all varieties.

In contrast, displacement is a very different mechanism to transference, and results from the coherent inter-action between the electric and magnetic fields of induction where they are in phase both spatially and temporally, a condition that is never possible with transference and not normally observed within electrical circuit measurements. Accordingly displacement is a phenomenon where the electric and magnetic fields of induction cannot be distinguished from the other electrically, they are essentially undifferentiated, both are acting in the system and acting together as one induced field. This yields the very important property that the extent of the action does not vary with distance (space) and hence between source and load is a displacement of electric power rather than a transfer of electric power. When power is displaced that available at the source is also available at the load and at any point within the circuit connecting them. Displacement also leads to regeneration of the electrical system when source and load are correctly connected and the purpose or “need” of the circuit is established and maintained in a state of dynamic equilibrium.

Because the electric and magnetic fields of induction are spatially in-phase or coherent, measurement via voltages and currents does not appear easily possible, putting the phenomenon and displacement events outside the range of common electrical measurement equipment. What does appear to be observable is the impact that displacement has on form within a circuit, which includes:

1. Compression of oil in a tube.

2. Light from a bulb without radiated heat.

2. Attractive and repulsive forces on conductive materials.

3. Orthogonal streamers within an electric discharge.

4. Charging of capacitors and loads from “radiated” energy.

5. Charging of capacitors and regenerative properties in non-linear electrical systems.

6. Distribution of electric power without loss between multiple loads and a source.

7. Regeneration of otherwise expended electrical storage systems.

8. Telluric distribution of electric power without loss.

9. Telluric generation of electric power.

10. Generation of additional energy within a system.

In all these forms of unusual electric phenomena displacement appears to be a deeper driving mechanism. This mechanism appears always present in any electrical system or circuit, yet hidden behind the more basic mechanism of transference. Only when the need established dynamically in the circuit cannot be met (balanced) through transference, is the mechanism of displacement directly observable. In order to attempt to observe and find a way to measure and characterise displacement I have found it necessary to explore non-linear events within electric circuits, that is, starting with a circuit in steady state equilibrium and then unbalancing the fields of induction to such a degree that they cannot immediately be balanced through transference. In this state the effects of displacement can be readily observed along with the subsequent changes electrically in the circuit when transference catches-up with the initial displacement.

Any electric system that is exposed to repetitive non-linear events will show the effects of displacement albeit in low tension cases so small as to easily pass undetected, e.g. when a steady current has been established in the primary inductor of a transformer and is then interrupted, leading to the collapse of the magnetic field and a return of the stored energy, and with the assistance of displacement a higher than expected induced emf in the secondary of the transformer. However when the tension of the system increases it becomes much easier to observe the effects of the displacement mechanism, and hence experimental arrangements that introduce non-linear events in otherwise high tension balanced power transfer systems are very suitable for the exploration into the difference between the mechanisms of displacement and transference. Switched (impulse) systems appear to lead to unusual electrical phenomena that are the result of the displacement mechanism being exposed in the process of rebalancing the system dynamics and before transference takes over as the secondary mechanism of establishing the steady state, (transference being referred to as the primary state in our current understanding of electricity). In addition, any electrical system where transference can be “held-of” from establishing the steady state, will manifest and display the unusual electrical phenomena that result from the displacement mechanism.

An example of this concept relates to Tesla’s account of observing the closing of the main switch between a high tension DC dynamo and the parallel railway tracks with a distant load. In this case the purpose and hence electrical characteristics of the circuit are already established, however the pressure of electrification at the dynamo cannot establish the steady state electric power transfer immediately within the electrical system. In this case the process of transference of the differentiated electric and magnetic fields of induction cannot propagate round the circuit with sufficient velocity, leading to a condition where the purpose of the circuit is in an “invalidated” or transient state. In this case the mechanism of displacement must initiate and be called-forth, establishing the fields of induction into the proper and required states, and so leading to the observable manifestation of orthogonal, filament-like streamers, extending into the railway tracks for a brief moment as the primary mechanism of electric power balance, whereupon transference takes over yielding the known and measurable characteristics of electric power distribution through a parallel wire transmission line. It is by virtue of the enormous pressure of the DC dynamo, and the non-linear event of closing the main switch connection between the two, that in this case reveals the process of displacement so clearly to the observer.

In summary, for this introduction on the concepts of displacement and transference, displacement is a coherent phenomenon and mechanism where the electric and magnetic fields of induction are in phase spatially and temporally, and are effectively unified to one overall induction field. It is ever-present at a deeper level within electricity guiding the manifestation of electrical properties towards the purpose required of the circuit, medium, and boundary conditions presented to it. The mechanism of displacement is revealed in action when the continuity of the need of the circuit to re-balance to the steady state is disrupted or held-of, and cannot in the moment be addressed by the process of transference. In this case the mechanism of displacement is called-forth, and whose action on the form can be observed, and is usually characterised by an injection of additional energy required to initiate the re-balance (speed-up) the process of transference. In this way displacement “moves” the now differentiated electric and magnetic fields of induction to the correct spatial and temporal synchronisation to allow transference to establish the final steady state electric power transfer according to the circuit, medium, and boundary properties. In turn this leads to the necessary changes in voltages and currents throughout the circuit and medium which can be readily measured with normal laboratory equipment. It could be seen that the mechanism of displacement relates to the principle of electricity, whereas the mechanism of transference relates to the properties of electric power.

In final conclusion to this part, displacement and transference are guiding principles, and also mechanisms, that explored and understood can show how our electrical machines, apparatus, and experiments can co-operate with the fundamental wheelwork of nature, and in so doing harness those underlying principles that lead to a more balanced and unified approach to the greater understanding of electricity, and in turn to the application of electric power to do work.

1. Heaviside O., Electromagnetic Theory – Volume 1, “The Electrician” Printing and Publishing Company Limited, 1893.

2. Dollard, E., Four Quadrant Representation of Electricity, A&P Electronic Media, 2013.

3. Dollard, E., A Common Language for Electrical Engineering – Lone Pine Writings, A&P Electronic Media, 2015.

4. Dollard, E. and Energetic Forum Members, Energetic Forum, 2008 onwards.

Generators, High Voltage Supply

High Voltage Supply

26th July 2018

The high voltage (HV) supply was one of the first items to be designed and constructed, and has subsequently been modified quite a few times to become what is now a flexible and reliable source of high voltage and current at the line frequency of 50Hz (UK standard), and up to sustained power outputs of 1600W, and peak power outputs up to 2500W.

Note: A high voltage supply is capable of delivering voltages and currents, even at lower powers, that are instantly lethal, and that any design and operation of a high voltage unit should be undertaken with great care by a trained and experienced individual. In my own case I was trained to work with high voltage equipment early in my career as an electronic engineer, and hence have opted for experimental flexibility, and maximum configuration, a power supply build that is accessible, open, and where high voltages could externally be exposed to the operator at certain key points. Careful design, implementation, and operation of such a supply is for the full responsibility of the individual concerned.

In the early days of this research it was unclear to me where in the experimental circuit interesting and unusual electrical phenomena originated from, whether it was the product of the generator, the tuning and driving units, the experimental coils themselves, the driven loads, the surrounding environment, or a combination of these factors. Later in the research I discovered that the generation of particularly displacement related events required a number of pre-conditions to be established, which involved the balance of the electric and magnetic fields of induction, which in turn involves all of the above factors, combined with a non-linear trigger, and with a defined load or “need” that cannot be met through the process of transference. These pre-conditions and the details pertaining to the generation of a displacement event will be considered and written-up in subsequent posts.

It was considered central to the early research, in replicating the key measurements and observations of other significant works e.g. Dollard et al^[1], that the generator design be as close as possible to those used, and especially considering that actual units and components may not be easily obtainable e.g. an original H.G. Fischer diathermy unit. However for this unit certain videos, internal pictures, and schematics where obtainable online and formed the basis of the first stage of building a suitable generator using easily obtainable parts and components. The overall generator to be used for experimentation is a combination of the HV supply detailed in this post, and driving a range of subsequent generator stages that transform the supplied HV AC voltage and currents at the line frequency, into higher frequency ac, oscillations, impulses, bursts, modulated waveforms, and other such driving waveforms as may be useful to the study of the displacement and transference of electric power.

Figures 1 show the HV supply which currently drives the different types of generator stages:

Fig. 1.1. High voltage base unit showing transformers, control circuits, and various different outlets (high voltage, mains, and low voltage).
Fig. 1.2. Three microwave oven transformers (MOTs) can be used individually, or connected in series, or in parallel.
Fig. 1.3. The SCR power controller for the MOTs has been modified to use a long-wire remote control for on-off and power level.
Fig. 1.4. High voltage transformers are cooled when required by twin cooling fans, and a mains bulb indicates power supplied to the primary of the transformers.
Fig. 1.5. The MOTs are individually switched on-off. The left two are connected in series via their metal core, whilst the right one is isolated from its metal core.
Fig. 1.6. All transformers are mounted on a large metal heatsink, and fan cooling is required for long operating runs at output powers > 500W.
Fig. 1.7. AC mains input power is passed through circuit breakers, filtered, and then fed to SCR power controller.
Fig. 1.8. The remote control allows distance operation of on-off and power adjust from zero to maximum, whilst working directly with the experiment at hand.
Fig. 1.9. The MOT is isolated from its metal core by detaching the core connected end of the secondary, and then routing the connection via high voltage cable to the output connectors.
Fig. 1.10. The two left hand metal core connected MOTs have their primaries connected in anti-phase to allow series connection of the secondaries.
Fig. 1.11. High power load resistor 100Ω @ 100W provides a stabalising load for the transformers when driving high impedance experimental loads.
Fig. 1.12. The high voltage supply with mounted upper deck to support generator apparatus.

The circuit diagram for the HV supply is shown in Figure 2 below, or click here to view the high resolution version. The schematic should be referenced for the subsequent circuit description:

Fig. 2.1 High voltage supply schematic showing the supply input stages, remote control, and the transformer output stages.

AC line power at UK standard 240V 50Hz is fed via a high current (16A) 3-pin connector to a domestic distribution box with 3 circuit breakers at 6A, 10A, and 20A. The 6A circuit breaker feeds a low voltage switched-mode power supply unit providing 15V @ 3A and is used to power low voltage circuits in the HV supply and the generator stage including, pre-amplifiers, fans, control electronics, measurement devices, indicators, meters, and any other low voltage units. The HV supply is arranged with a number of low voltage two-pin power jack sockets to supply low voltage generator stage requirements on the upper level. The 10A circuit breaker powers filament transformers for vacuum tube generator stages, and auxiliary devices requiring line voltage AC, via a suitable mains output connector in the form of shielded 3-pin connector, and ceramic connection block for ad-hoc connections. The 20A circuit breaker feeds the high voltage transformers via a suitable power controller.

The distribution unit also has space for an incoming line RCD breaker, but has subsequently been removed, as it was found to be too sensitive to some experiments where power is reflected back into the HV supply, and causing the RCD to cut the power during the experiment. As an alternative a larger RCD was incorporated into the mains distribution for the laboratory, and separate mains circuits fed via a UPS (uninterruptible power supply) to measurement and test equipment, and computers. This arrangement prevents sensitive equipment from inadvertently being switched-off and/or rebooted during certain experiments when the lab RCD would disengage to protect the input supply. Having the test equipment running under these circumstances has proved key to understanding conditions and events within the experiment that have caused large reflections back through the mains supply.

The 20A circuit is first passed through a high current line filter which is used to prevent higher frequency electrical disturbances from being reflected back into the mains supply, and offer a measure of isolation between the two. The output of the line filter is fed to a power controller which enables the variable control of power supplied to the high voltage transformers. This HV supply was specifically designed around the use of the microwave oven transformer (MOT) as the high voltage part of the supply. MOTs are very readily available, and have proved to be a strong and robust transformer for this type of supply. The transformers used are all Galanz GAL-900E based which nominally produce 2100V_RMS @ 900W ~0.45A_RMS, and are quite common in UK domestic microwave ovens. The MOT represents a significant inductive load to the incoming AC supply, and uncorrected will reduce the power factor from the ideal 1 to ~0.6. To correct for this and reduce the draw on the incoming supply a power factor correction (PFC) capacitor can be used at the AC line input (after the line filter and before the SCR). A 20µF AC PFC capacitor can be used in order to correct for a single MOT. For 2-3 MOTs being used together this can be increased to 40µF.

The MOT is a transformer designed to drive a specific impedance load, (magnetron via a voltage doubler and tuned with a series capacitor), with the minimum quantity, and hence cost, of copper, and with the cheapest and simplest manufacture methods and components. This leads to certain drawbacks in the transformer characteristics, and most especially saturation of the transformer core when driven open-circuit, or connected to a higher than intended load impedance. The core is cheaply manufactured from steel laminate and then welded together which shorts the laminates out, greatly increasing the core saturation rate when adequate power is not drawn from the transformer. A detailed study of the characteristics of the MOT has been presented by Wokoun^[2].

The easy core saturation requires the current to be restricted in the primary coil. This is not easily done directly with a variac, as is usual for variable output control of a transformer, since the core easily saturates at low input primary voltages leading to large run-away currents in the primary, rapid core heating, and ultimately destruction of the transformer from excessive heating, not to mention the dangerous risk of a transformer fire which is very hard to deal with due to extreme heating of the steel core even after the power has been cut-off at the input. Instead the current must be restricted either via an inductive load in the primary circuit, or much better, an SCR power controller.

A suitable simple series inductor is the primary of another MOT, (with the secondary shorted), connected in series with the primary of the MOT to act as the HV transformer. Alternatively the secondary of another MOT, (with the primary shorted), can be connected in series with the secondary of the HV transformer MOT, but more heating tends to occur in this configuration from the higher secondary impedance. The series primary connected MOT was found to limit the output current to a degree, and made adjustment with an input variac possible, with less chance of core saturation, but with however limited overall range of adjustment and suitability to changing load impedance. The advantage however of this first method is that the output of the MOT (not in core saturation) is a complete sine wave. In core saturation the output becomes progressively distorted towards a heavily clipped sine wave. It was concluded that this method of power control would be too limited for the wide range of generators that the HV supply would be driving.

The second and preferred method is the SCR based power controller, (similar to a light-dimmer controller but more powerful), which controls the on part of the sinusoidal cycle, and hence controls the overall power delivered to the transformer, which effectively restricts the core saturation whilst providing variable control of the output power. Suitable SCRs are very easily and cheaply available, and a complete unit with an output power of up to 3kW has been used in the HV supply. The disadvantage of the SCR is that the output is no longer a sine wave, but rather a distorted waveform that represents a small part of the total cycle. This has however provided some unexpected benefit in burst and impulse modes that will be discussed in the generator posts, but suffice to say here that the fast SCR turn-off can create very large voltage spikes in the MOT primary as the field collapses, which in turn produces strong impulses in the secondary at the line frequency. These impulses in the secondary, when fed directly to the experiment without a capacitor tank circuit, acts as one method of generating a non-linear trigger for a displacement event.

When working directly with the experiment at hand it is not convenient to keep walking backwards and forwards to the power supply to adjust power level or turn on or off the supply. To enable more distant control of the SCR the variable resistor used to control the power level was removed from the SCR circuit, and positioned in a small plastic box along with an on-off switch. The control box was then attached to the SCR via a long two-wire mains lead (5m), where on-off function is created by switching a higher resistance into the two-wire line, and hence holding the SCR in the off condition, which is also the case if the remote box is disconnected from the SCR power controller. Power control is affected from the variable resistor by reducing the resistance from 500kΩ down to 0Ω, which progressively turns the SCR on for a proportion of the ac line cycle.

Figures 3 below show waveforms from the HV supply at a range of different points in the high voltage supply, and including the high voltage rectifier and tank capacitor at the output to form a load.

Fig. 3.1 Line input waveform showing badly clipped sinusoidal waveform at ~ 240V_RMS and 50Hz
(UK standard).
Fig. 3.2 Input (1) and output (2) waveforms of the SCR power controller at 25%, and disconnected from the MOT input.
Fig. 3.3 SCR at 50%.
Fig. 3.4 SCR at 75%.
Fig. 3.2 SCR at 100%.
Fig. 3.6 Output of the SCR when connected to a MOT driving a high voltage rectifier and capacitor, and with the SCR set at 75%.
Fig. 3.7 Output of the MOT when driving a high voltage rectifier and capacitor, and with the SCR set at 75% output.
Fig. 3.8 High voltage output from the rectifier and tank capacitor, showing a constant dc voltage at ~3kV.

The output of the SCR power controller is passed through a system of connections to allow the MOTs to either be driven directly from an external source as required, or by direct connection to the SCR. Each individual MOT can be switched independently to the SCR output allowing the transformers to be used individually or combined in parallel or series combinations to increase the available output current or voltage. The output of the SCR is also fed to a 25W mains incandescent lamp which indicates clearly to the operator when voltage is applied to the input of the one or more of the MOTs. This is a simple but important safety factor when working in the experimental environment, and is a rapid but not exhaustive check to the running status of the high voltage supply. It must also always be remembered that considerable energy can be stored in the generator components, such as tank capacitors etc., and that a no visible lamp output is not a direct indication that it is safe to touch any part of the high voltage circuits prior to the appropriate discharge procedures.

The MOT is a cheaply manufactured component with minimum materials and quality, and hence the high voltage winding isolation to the steel core will not usually withstand voltages in excess of ~1.5 times the nominal designed output. This makes it difficult to combine MOTs in series where the core connected terminal of the secondary has been detached from the core in order to float the secondary, whilst keeping the steel core connected to earth for safety purposes. In this configuration the open-circuit peak voltage of the secondary can reach almost ~6kV from 2 series connected MOTs, which can easily arc-over to the steel core through the secondary insulation. When allowed to happen for any period of time the secondary coil is easily permanently damaged.

To overcome this problem and to enable two MOTs to be connected safely in series (both cores earthed), the MOTs are connected in series anti-phase, or center-tapped arrangement. In this configuration the two cores are connected together to earth, which also means the two core connected ends of the two secondary coils are also connected to earth. The primaries of the two transformers are then connected in reverse phase to each other, (as shown in the circuit diagram), such that one transformer produces +V_HT out, and the other transformer produces -V_HT out. The total output voltage of the series connected secondaries is 2V_HT, and the maximum secondary to core voltage on either transformer is only V_HT, preventing any secondary to core breakdown.

Of the three available MOTs in the high voltage supply, two are centre-tap connected, and one is floated from the core. This combination was found to be most flexible where the centre-tapped pair are suitable for driving spark gap based generators, and the floated individual is most suitable for driving vacuum tube based generators and if required in conjunction with a diode voltage doubler. In some generator configurations it was necessary to reduce the secondary current using a power resistor, which also in some specific cases helps to stabilise changes in power factor when driving varying or fluctuating high impedance loads from the generator outputs. When and where required a fan-cooled 100Ω 100W wire-wound resistor was used to reduce secondary currents and stabilise the supply output impedance to the next stage. For sustained outputs the MOTs and output components are cooled using a pair of low voltage fans which are manually switched as required.

Overall the high voltage supply has proved to be robust and versatile in providing high voltage in a variety of configurations to a range of different types of generator circuits. The design of the high voltage supply makes it easy to use in the experiments, with accurate and remote control of the output, and constructed with basic and readily available components.

Click here to continue to the 811A vacuum tube generator.

1. Dollard, E. & Lindemann, P. & Brown, T., Tesla’s Longitudinal Electricity, Borderland Sciences Video, 1987.

2. Wokoun, P., Investigations on Using a Salvaged Microwave Oven Transformer, 2003, KH6GRT Website

Experiments, Flat Coil, Measurements, Single Wire Currents

Single Wire Currents

15th June 2018

Part 1 of single wire currents investigates the voltages and currents generated in the secondary coil, and connected load circuit, when the primary is driven from a suitable generator. In this part the generator used is a high voltage vacuum tube oscillator which derives the feedback for oscillation directly from the dominant flat coil resonant frequency.

The design, construction, and measurement of this generator, and its matching and tuning circuit, will be reported in subsequent posts. For clarity here the following different types of generator have been built and tested in a wide range of different experiments:

1. Vacuum tube generator driven either by an external high power oscillator, or directly as a self-tuned oscillator using feedback from the secondary coil. Can be driven in CW (carrier or continuous wave), burst, or modulated modes.

2. Spark gap generator, (static or rotary), driving directly a primary matching and tuning circuit, (tuning circuit as shown in Fig. 1.4 below).

3. Spark gaps driving a modern replica of an H.G. Fischer diathermy generator.

4. An original 1920’s H.G. Fischer diathermy generator.

Experiments in single wire currents investigate the interesting and unusual properties that result from high voltage and often high frequency waves emitted from a suitable source or generator and guided by a single wire to a load. The single wire nature means that power is passed from the generator to the load, and where the load is able to utilise this power to do work, through only a single wire. In a standard electric circuit a source of electric power such as a battery or an oscillator would be connected from both the +ve and -ve terminals for a current (dc or ac) to move around the circuit, and doing work in the circuit dependent on the characteristics and nature of the circuit. In this case if one of the terminals were removed, the circuit would be considered open-circuit, no current would flow, and no power could be utilised to do work within the circuit. In the single wire case the power conveyed through the electric and magnetic fields of induction easily do measurable work e.g. lighting an incandescent bulb, whilst the current in the circuit appears to be guided only by a single wire, that is, there is no obvious return wire for the current to pass back to the generator and create the required “circuit” for the classical conduction of electric current.

In part 1 of this experiment a vacuum tube generator is used to apply an rf sinusoidal (ac) current to the primary of the flat coil in CW mode. By extension of the magnetic field of induction to the secondary coil a magnified electric field of induction (emf) is induced across the secondary of the coil. When the secondary coil is further connected to a load via a wire at the bottom-end, or outer-end, an oscillating current (resulting from a reciprocal inter-action between the electric and magnetic fields of induction) is guided by the conductor of the wire to the load. In conjunction a pick-up coil is used behind the secondary to induce a small part of the magnified wave and feed this back to the vacuum tube oscillator. This positive feedback signal drives the oscillator at the dominant (tuned) frequency of the flat coil, in this case the lower resonant frequency F_L at ~ 1850kc/s where C_P ~ 900pF. In this way the circuit can be measured at a single frequency which can be tuned and adjusted using the primary capacitance C_P.

Figures 1. show the generator connected flat coil 1S-3P to be used in the single wire current experiments, and including the primary tuning circuit with primary capacitance C_P, in this case a 4kV vacuum capacitor:

Fig. 1.1. Flat coil 1S-3P used as the generator coil for experiments in single wire currents.
Fig. 1.2. Flat coil 1S-3P constructed with PTFE coated multi-stranded wire, and copper strip primary.
Fig. 1.3. Rear side of the flat coil showing generator feed and matching unit, and secondary pick-up coil to drive vacuum tube feedback.
Fig. 1.4. Generator feed and matching unit consisting of the primary capacitor C_P, and feed capacitors and 1B22 spark gap tubes for use with a spark gap generator.
Fig. 1.5. Generator feed unit is attached to the flat coil with copper strip and the primary capacitor is attached with silicone micro-stranded wire at the correct weight of copper connection point.

Figures 2. show the single wire current experimental apparatus, including measurement equipment and probes:

Fig. 2.1. Single wire currents experimental setup showing the vacuum tube generator, flat coil, load, probes, and test equipment.
Fig. 2.2. The flat coil connected to the load bulbs via an RF current meter, and measuring secondary voltage and current. The output of the load is connected only to a single insulated wire.
Fig. 2.3. Final load bulbs do not light when the output lead is disconnected.
Fig. 2.4. Aluminium leaf attracted towards the bulb load, and held in place whilst the generator is switched on.
Fig. 2.5. Aluminium leaf attracted to the bulb load, a source of light and rf emitter.
Fig. 2.6. Rear side of the generator coil showing matching unit connected to the generator, and rf current meters in the primary and secondary circuits.
Fig. 2.7. Vacuum tube generator delivering 479W at 1850kc/s to the generator matching unit. Voltage and currents in the primary and secondary circuits are being measured using an oscilloscope.
Fig. 2.8. Radiated power from the experiment is picked up by a small whip antenna connected directly to a spectrum analyser.
Fig. 2.9. With the load output wire disconnected the fundamental frequency has shifted to 1860kc/s, with no current measured in the output wire (red osciloscope trace).

To view the large images in a new window whilst reading the explanations click on the figure numbers below:

Fig 2.1. Shows the overall experimental apparatus, measurement probes, and equipment. The vacuum tube generator feeds the connections to the tuning unit with the primary capacitance. A high voltage differential probe Pintech DP-50 is connected across the primary capacitance to show the electric potential V_P applied across its terminals. A current probe Tektronix A6303 is connected around the wire between the primary capacitor and the plates of the vacuum tubes to show the electric current I_P moving through the primary circuit. Inserted between the high voltage tank capacitor and the input to the primary is a Weston model 425 rf ammeter (either 1A or 5A full scale deflection (fsd) dependent on generator output, and with internal thermocouple), to additionally monitor the primary rf currents I_PRF.

In the secondary circuit the top-end of the flat coil is terminated with a 240V 5W (UK standard) neon bulb to act as an indicator of the magnitude of induced electric potential or tension, and to contain the top-end with a defined impedance. This containment assists in stabilising the resonant cavity formed by the secondary coil, and without significantly loading the coil and effecting the upper and lower resonant frequencies, or the Q-factor. The bottom-end of the secondary coil is connected by short wire to another Weston model 425 rf ammeter (250mA fsd) combined with a parallel 5Ω shunt to make 500mA fsd and to monitor the secondary rf currents I_SRF.

The bottom-end of the coil is also connected to a high-voltage probe Pintech HVP40 40kV 1000:1 passive probe to monitor the secondary potential V_S at the lower terminal. The output of the secondary ammeter is connected to the load, which in this case is 4 x 240V 25W (UK standard) pygmy bulbs with vertically laced filaments. The bulbs can be connected in a variety of arrangements, but were here used in a two parallel twin series connected arrangement so that all 4 bulbs will light as the load. The output of the load was connected to an 80cm flying lead. Secondary current I_S was monitored in various places using a second Tektronix A6303 current probe.

The outputs of probes V_P and I_P from the primary, and V_S and I_S from the secondary, were passed to the inputs a four input oscilloscope HP54542C for measurement and comparison. In addition the signal V_P was fed to a Tektronix DC5009 Universal Counter to confirm the oscillation frequency of the primary circuit. This frequency of oscillation was also monitored via a Tektronix 7L5 spectrum analyser fed by a small whip antenna at the input. Throughout the experiment the Tektronix current probes 2 x A6303 connected to AM503B current probe amplifiers were set on 1A AC /division. The total input power to generator P_IN, (input to the high voltage transformers only), was monitored using a Yokogawa WT200 digital power meter.

Fig 2.2. Shows that at an input power of P_IN = 319W @ 1851kc/s, I_PRF~ 700mA, I_SRF ~ 240mA (2 x 120), and a 80cm fly lead connected to the output of the load bulbs, that all the bulbs are lit with the first two bulbs being lit brightly whilst the second two bulbs are only dimly lit. The measured waveforms will be considered in more detail in Figures 3.

Fig 2.3. Shows that under the same electrical conditions with the fly lead removed from the second load bulbs the intensity of the bulbs is greatly reduced. The first set of load bulbs are now dimly lit, whereas the second set of load bulbs are not visibly illuminated. I_SRF has also reduced considerably to ~ 100mA (2 x 50mA), whilst I_PRF increased slightly to ~ 770mA, at a P_IN = 318W @ 1860kc/s. Here the frequency of oscillation has increased slightly due to the reduction in wire length with the fly lead removed, although vacuum tube generator has compensated automatically to shift resonance to the new resonant frequency via the secondary pick-up coil. The most important feature here is that in single wire current experiments loads will not power when no fly lead or terminating lead is connected to their output. In the case of a bulb it will not light when it is the last device connected to the single wire.

Fig 2.4. Shows the effect of introducing a conductive material close to the load in this case an aluminium leaf suspended by masking tape from an insulated support. Within a certain distance the aluminium leaf is attracted to the bulb outer glass surface and can remain held in this place until the generator is turned off. It appears a force is applied to the aluminium leaf that will move and/or retain the leaf in a distance offset from the vertical. This unusual result has been investigated in a variety of different ways and will be introduced here, to be further investigated and described in subsequent parts.

In the case of the CW vacuum tube generator (VTG-CW) the waveform induced in the secondary circuit is a steady and constant oscillation at a single frequency. This is a very linear and determinate condition and has been found to have the least intensity on the phenomena of attraction of conductive materials. At input powers typically 250W upwards in the experimental apparatus shown the aluminium leaf is very slightly attracted to the bulb glass. If placed only 1mm from the surface then the leaf will be pulled directly from vertical to a point on the glass bulb surface and held there. For distances x between the leaf and the bulb in the range 1mm < x < 15mm, and for the VTG in CW mode, the leaf can be held in place when initially placed on the bulb surface. Above ~15mm the aluminium leaf will not be retained on the bulb surface but will swing back to the vertical position.

The magnitude of the force applied to the aluminium leaf increases with the input power P_INto the generator and hence I_SRF in the secondary wire. The overall effect is similar to observing a magnetic metal attracted to a magnet at close range, or the effect of electrostatic attraction in the case of opposite charged metal plates spaced slightly apart. In this case however it appears that the effect is based on the electric field of induction being dominant in the scenario rather than magnetic field of induction. When a permanent magnet is introduced into the experiment it has no influence on the attraction of the aluminium leaf either in being attracted towards the bulb, away from it, or being held on the bulb surface.

The intensity of the attraction and hence the magnitude of the applied force on the leaf has been found to increase significantly with burst, impulse, and modulated waveforms. With a burst or impulse waveform from the generator it is easily seen that at P_IN > 400W the leaf can be instantly attracted to the bulb and move from the vertical over distances as much as 20mm, and then held there strongly on the surface of the glass. in this case even with the generator turned off the leaf can be retained for up to 60 seconds on the surface of the bulb before being released and swinging back to the horizontal.

Other types of leaf material have also been tested, and those found to readily be attracted and retained to the bulb glass have a conductive element to them, including metals like aluminium and copper, organic materials such as living tissue, plant matter (e.g. leaves), and paper, cardboard, and woods with a certain content of moisture in them. In the case of organic living tissue the presence of my hand in the vicinity of the light bulb, but not touching, greatly increases the effect even in CW mode. For man-made synthetic materials such as plastic and other insulating mediums there is normally no discernible attraction towards the bulb. At very high voltages and high input powers P_IN > 1000W a plastic leaf was found to attracted to the bulb surface over a tiny distance < 0.5mm but could not be retained on the surface of the bulb even when placed directly on the surface.

With the aluminium leaf the voltage on the leaf was measured during the process of attraction and was found to rise to a high dc potential usually in the order of several hundred volts in the experiment thus described. This indicates a form of “charging” like the plate of a capacitor when exposed to a dc potential higher or lower than the surrounding environment. In this case the electric field of induction appears to have created a region of potential difference and tension between the material of the leaf, where the leaf has become “charged” to an opposite polarity than that present on the glass surface of the bulb. It is conjectured here that an electric wavefront (a positive dc level or impulse rather than a varying sinusoid) is emitted from the exposed wire of the bulb filament (itself a tiny extra coil and leading to an imbalance between the magnetic and electric fields of induction). These continuous wavefronts result in charge accumulation on the surface of the conductive material which establishes an electric field between the bulb filament and the conductive material. The electric field results in a force exerted on the aluminium leaf which is pulled towards the glass surface. As the conductors of the filament and the leaf are prevented to come into contact by the glass bulb the electric field is not collapsed by shorting the two together, and the leaf can be retained firmly on the glass surface as it remains “charged” by the presented wavefronts.

It is suggested that the attraction is not likely to be magnetic in nature, and as a result of eddy currents in the conductive material induced by the presence of a time varying magnetic field, as the phenomena cannot be influenced by other magnetic fields in very close vicinity, such as permanent magnets and electromagnets. It would be expected that the magnetic field generated by eddy currents in the leaf would be disturbed by the introduction of a strong permanent magnet, however no such disturbances have been observed or measured.

To eliminate effects due to convection and movement of air due to heating of the glass bulb a control experiment connected the same bulb type, a 240V 25W pygmy bulb, to a normal domestic ac outlet so that it would light to normal intensity and heating. The aluminium leaf was then placed in very close proximity to the bulb surface ~ 0.5mm with no discernible movement towards the bulb over any length of time the control experiment was conducted.

Fig 2.5. Shows in close-up detail the attraction of an aluminium leaf to the surface of the load bulb and being retained on the surface until the generator is turned off. In this case with the VTG in CW mode the attraction is not strong enough to pull the leaf from the vertical over a distance of 15mm to the bulb surface. The applied force is however strong enough to retain the leaf on the surface of the bulb at a distance of 15mm from the vertical, and once placed on the surface of the bulb.

Fig 2.6. Shows the experimental apparatus from the reverse side with the generator attached to the tuning unit, the rf ammeters in the primary and secondary, and the generator tank capacitor meter in the far bottom right showing a tank voltage of ~ 800V dc.

Fig 2.7. Shows the vacuum tube generator, primary measurement probes in the background, and the test equipment setup with P_IN = 479W, the primary and secondary voltages and currents measured on the oscilloscope, and the measured oscillation frequency of the primary F_P = 1.850Mc/s on the frequency counter.

Fig 2.8. Shows the spectral response of the emitted electric field in vicinity of the experimental setup and as measured by the Tektronix 7L5 spectrum analyser connected to a small whip antenna as shown in the bottom right of the picture. The spectral response shows a significant peak at ~1850kHz, and small possibly “artefact” peak at ~1950kHz.

Fig 2.9. Shows particularly the change in oscillation frequency measured in the primary circuit when the fly lead was removed from the output of the bulb load. The oscillation frequency of the experiment changes from ~1850kc/s to ~1860kc/s.

Figures 3. show the voltage and current waveforms for the primary and secondary and their phase relationship:

Fig. 3.1. Oscilloscope traces for primary (1,2) and secondary (3,4) voltages and currents. The output lead is connected to the load, all bulbs are lit.
Fig. 3.2. Oscilloscope traces for primary (1,2) and secondary (3,4) voltages and currents. The output lead is disconnected from the load, final bulbs are not lit, and secondary current is monitored in the disconnected load wire.
Fig. 3.3. Spectrum analyser signal picked-up using a small whip antenna positioned 3m from the experiment, and showing the resonant frequency at ~1850kc/s.

Fig 3.1. Shows the primary and secondary voltage and current measurements V_P (trace 1) and I_P (trace 2), and V_S (trace 3) and I_S (trace 4) respectively. V_P is a sinusoidal oscillating voltage V_PK-PK ~ 2kV. I_Pis more in the form of a pulsed current where the trace is calibrated 1V per amp and showing I_PK-PK of ~ 2A. The phase of the current I_Pis leading V_Pby ~90° indicating that the generator appears to be driving a reactive load that is predominantly capacitive in a class-C amplifier arrangement. This is to be expected as the 180° phase change of the primary has been shown to exist at a much higher frequency than the impedance maximum for the primary would indicate. Operated in this way the primary and secondary are not at resonance simultaneously, the primary circuit is oscillating with a driven ac, whilst the secondary is acting as a free resonator at its tuned resonant frequency which determines the driven frequency in the primary.

As the voltage V_P rises across the primary the current I_P is maximum and falls rapidly as the primary capacitor C_Pis charged by the tank capacitor, on which that energy is released through the inductance of the primary coil reversing the current flow and discharging C_P. This yields current pulses of sufficient magnitude for the magnetic field of induction to dominate and extend to the secondary coil. The secondary coil is not tightly coupled to the primary and so can reasonably resonate freely as the generator oscillates at a frequency determined by feedback from the secondary to the generator pick-up coil.

Using the VTG in cw mode it is important to note that the secondary is constantly being excited by the primary in a linear continuous fashion. There is no charge and discharge phase in the secondary as would occur in a burst or impulse driven primary. In this case the VTG is driving the flat coil in a very linear condition where the system operates at one set frequency, and the dominant majority of energy is conveyed at the fundamental resonant frequency, with very little contribution from harmonics. In this case we would expect phenomena that arise from the imbalance between the electric and magnetic fields of induction to be minimal, which is so far confirmed by measurement of single wire phenomena including deflection of conductive materials, and dc charging of capacitive loads.

The freely resonating secondary shows V_Pand I_Pwhich are in phase in traces 3 and 4, which is to be expected for a freely resonating coil driven with a very linear continuous wave. V_S at the bottom-end or outer-end of the secondary coil is ~1kV_PK-PK, and the current I_P measured by the current probe prior to the load (as shown in Fig. 2.2) is ~ 2A_PK-PK (1V per amp calibrated on the current probe amplifier).

Fig 3.2. Shows the change in waveforms when the fly lead is removed from the end of the load, and the secondary current probe is connected through the fly lead. The frequency of oscillation has increased due to the reduced wire length in the experiment to ~1860kc/s (as measured by the frequency counter and spectrum analyser, rather than the marker frequency of the oscilloscope). The primary waveforms V_Pand I_P remain largely the same in amplitude, phase, and form. The secondary voltage V_S has increased as the effective load is reduced in the secondary, and I_S has gone to zero as the fly lead, from which the current is being measured, has been disconnected from the output of the load. In this case the final load bulbs were not lighted, and the first load bulbs were lit only dimly with a significant reduction in I_SRF.

Fig 3.3. Confirms the electric field detected in the vicinity of the experiment throughout the measurement period, where the pick-up whip antenna is located ~ 3m from the load bulbs.

To view the large images in a new window whilst reading the explanations click on the figure numbers below:

Figures 4. show the Z11 input impedance characteristics of the experimental apparatus:

Fig. 4.1. Z11 input impedance as seen by the generator of the experimental setup including the matching unit, flat coil 1S-3P, RF meters, load bulbs, and measuring probes, and the single output wire connected to the load.
Fig. 4.2. Complete experimental setup with the single output wire disconnected from the load.
Fig. 4.3. Experimental setup with the bottom-end of the flat coil disconnected from the wire feeding the secondary rf meter, (meter, load, and probes disconnected).
Fig. 4.4. Complete experimental setup with the primary capacitor disconnected.

To view the large images in a new window whilst reading the explanations click on the figure numbers below:

Fig 4.1. Shows the small signal input impedance Z11 as seen by the generator of the complete experimental apparatus with all measurement probes connected, and the fly lead connected at the output of the bulb load. The impedance characteristics show that the experiment tuning is operating very close to the balanced point between the lower and upper resonant frequency, F_Land F_U, of the flat coil. This is the point where there is expected to be best balance between the electric and magnetic fields of induction between the primary and the secondary coils, and in this case the best experimental starting point when investigating the displacement and transference of electric power through non-linear processes. F_Lmeasured when running the single wire current experiments was ~1850kc/s, and from the impedance characteristics 1889kc/s a variation of ~2%, and most likely due to differences between the small-signal and large-signal operation points of the flat coil, tuning components, and generator mode of operation (cw class-C).

Fig 4.2. Shows the result of removing the fly lead the length of wire in the secondary section of the experiment has been reduced, and hence the frequency increased from ~1850kc/s to ~1860kc/s. This is also indicated by the impedance characteristics where the 180° phase change frequency F_Ø180 has shifted from 2345kc/s in Fig. 4.1 up to 2388kc/s. This has also created a greater imbalance between F_Land F_U.

Fig 4.3. Shows the result of removing the experiment from the bottom-end or outer-end of the secondary coil. All frequencies are shifted up due to the change again in wire length, and also the change of impedance at the bottom-end from lower to higher, and away from the λ/4 mode.

Fig 4.4. With the primary capacitance C_P removed the impedance characteristics of the experiment revert to the loaded properties of the secondary coil with a single resonant frequency, and there is no established balance between the electric and magnetic fields of induction between the primary and the secondary.

Summary of the results and conclusions so far:

1. Single wire currents have been observed and measured using a flat coil driven by a vacuum tube generator in cw mode. The current measured in the single wire, and its properties thus far observed, would appear to suggest that rf energy from the wire is escaping along its length to the surrounding environment which acts as an energy sink, ground, or -ve terminal, which then effectively completes the circuit. High energy rf as a result of the magnified voltage produced by the secondary coil, is easily radiated from all parts of the conductor that forms the wire through to the end of the fly lead. With this being the case, and with the voltage and current being in phase in the secondary, real power is generated to drive the load bulbs which emit both light and heat. With the fly lead removed the final load bulbs do not light as there is insufficient length of conductor to act as a suitable radiator or sink “to ground”. It is expected that any load connected to the end of the single wire will not be driven as there is insufficient energy sink on the output of the load to enable a current to be developed through the load. With this being the case the energy sink is distributed along the length of the wire so that the current along the wire would not be a constant value, as might be expected normally for the current flowing through a circuit. In part 2 of single wire currents it will be necessary to measure the magnitude and phase of the current along the wire length as a function of distributed load which would then allow a more accurate picture, and hence interpretation, of single wire current action in a circuit.

2. Standing waves were not observed or measured along the length of the single wire in this experiment, but rather the magnitude of the oscillating voltage appears to remain relatively constant along the length of wire, whilst the current reduces with load and distance along the wire. This will be further investigated in part 2 where a more accurate voltage and current distribution will be measured with wire length and load distribution.

3. A force applied to a conductive medium in close proximity to a load on the wire, in this case a lighted incandescent bulb filament, has been observed and investigated at first stage. The phenomena, at this stage, appears to result from a form of electric attraction between the filament of the bulb the emitter, and the conductive medium. The effect does not appear to be influenced by other close proximity magnetic fields such as permanent magnets, and electromagnets, which also suggests that the phenomena does not result from eddy currents generated in the conductive medium. A range of different materials have been tested, and all that show a significant attraction towards the load bulb, have a conductive element or property. The effect is also greatly amplified in the presence of a significant energy sink such as the hand of a person. In cw mode no discernible force could be registered on the surface of the hand when placed in close proximity to a load bulb. This has been subsequently demonstrated when driving the generator in burst or impulse mode and will be presented in detail in subsequent parts.

4. The impedance characteristics indicate that the complete experiment was operated in a well-balanced mode of the flat coil, which suggests a good starting point for further, and more detailed investigation, of the displacement and transference of electric power through non-linear events.

Click here to continue to Transference of Electric Power – Part 1.

1. A & P Electronic Media, AMInnovations by Adrian Marsh, 2019, EMediaPress

2. Dollard, E. and Energetic Forum Members, Energetic Forum, 2008 onwards.